A Novel Current-Fed Boost Converter With Ripple ... - Semantic Scholar

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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 57, NO. 6, JUNE 2010

A Novel Current-Fed Boost Converter With Ripple Reduction for High-Voltage Conversion Applications Ching-Shan Leu, Member, IEEE, and Ming-Hui Li

Abstract—Employing a new rectifier circuit, i.e., a novel current-fed boost converter with ripple reduction, is proposed in this paper. It features high conversion ratio with smaller transformer turn ratio, recovery of transformer secondary leakage energy, low voltage stress on the rectifier diodes, and lower inputand output-current ripples with minimum component count. Therefore, high efficiency and power density can be achieved under high-frequency operation. Moreover, the new rectifier circuit can be applied to all current-fed power topologies for high-voltage conversion applications, such as fuel-cell-powered systems. The operating principle, theoretical analysis, and design considerations are presented. To demonstrate its feasibility, a 150-kHz, 16–22-V-input, and 200-V/400-W-output converter is implemented and tested. Index Terms—Boost, current-fed, low current ripple.

I. I NTRODUCTION

T

HE fuel cell (FC) is considered to be one of the promising alternative energy sources for the future [1]–[5]. However, it has several inherent limitations, such as wide-range low dc output voltage, slow dynamic response under load variations, and low fuel efficiency due to high ripple current. Therefore, the FC is generally integrated with the power-conditioning system (PCS). As shown in Fig. 1, the PCS consists of a dc–dc converter, a bidirectional dc–dc converter, a dc–ac inverter, and an energy storage. Among them, the dc–dc converter is the most difficult to design because it has to deal with problems caused by high-output-voltage and high-input-current operating conditions. To realize the high output voltage, generally, a high-turnratio transformer is required. A large leakage inductance and a large parasitic capacitance are thus induced, resulting in the generation of high voltage and current spikes on power devices. Therefore, a topology with a smaller turn-ratio transformer is highly recommended [5], [6]. To handle the high input current, on the other hand, a converter with a continuous input current is preferred. In addition to decreasing the conduction loss due to its smaller input-current

Fig. 1.

Block diagram of an FC power conversion system.

rms values, the number of input electrolytic capacitors can be minimized. The current-fed configuration is thus selected as the dc–dc converter topology in preference to the voltage-fed configuration. Several current-fed dc–dc step-up converters are proposed in the literature [7]–[26]. Although problems caused by parasitic components can be alleviated, a snubber circuit has to be used to suppress the voltage spike on rectifier diodes, which are rated with at least two times output voltage. Consequently, the converter’s efficiency is degraded. Moreover, a larger high-voltage-rating capacitor is also needed to alleviate the output-current ripple, and the power density is thus limited. To provide a comprehensive solution, a current-fed power converter having built-in voltage-clamping, lossless snubber, and output-current ripple reduction functions with minimum component count has to be explored. Therefore, a novel currentfed boost converter with ripple reduction (BCRR) topology is proposed in this paper [27]. Several features, such as high-voltage gain with smaller transformer turn ratio, recovery of transformer secondary leakage energy, low voltage stress on rectifier diodes, and low output-current ripple, can be obtained and make the proposed converter desirable for low-input-voltage to high-output-voltage applications, such as FC-powered systems. The theoretical analysis is verified by a 150-kHz, 16–22-V-input, and 200-V/400-W-output prototype. II. A NALYSIS AND C IRCUIT O PERATION

Manuscript received December 31, 2008; revised July 29, 2009 and October 16, 2009; accepted November 9, 2009. Date of publication March 1, 2010; date of current version May 12, 2010. C.-S. Leu is with the Department of Electrical Engineering, National Taiwan University of Science and Technology, Taipei 10673, Taiwan (e-mail: cleu@ mail.ntust.edu.tw). M.-H. Li was with National Taiwan University of Science and Technology, Taipei 10673, Taiwan. He is now with Skynet Electronics Company, Ltd., Taipei 11570, Taiwan (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIE.2010.2044114

The circuit diagram of the proposed BCRR is shown in Fig. 2. It comprises one input inductor (Lin), one transformer (T 1), two switches (Q1 and Q2), one clamping capacitor (C1), one output capacitor (Co), and two series-connected diode pairs (D1–D2 and D3–D4). The transformer has two primary windings (P 1–P 2) and two secondary windings (S1–S2) with a turn ratio of 1 : 1 : N : N. L1 and L2 represent two leakage inductances in the secondary. Without C1 and the connection between two diode pairs, it is identical to the current-fed push–pull converter with center tap (CF-PP-CT).

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LEU AND LI: A NOVEL CURRENT-FED BCRR FOR HIGH-VOLTAGE CONVERSION APPLICATIONS

Fig. 2.

Circuit diagram of the proposed BCRR.

Fig. 3.

Key waveforms of the BCRR.

To simplify the analysis of the proposed converter, all semiconductors are assumed to be ideal, and inductor Lin is assumed to be sufficiently large to be approximated by a current source with a value that is equal to the input current. The clamping capacitor (C1) and the output capacitor (Co) are assumed to be sufficiently large, in that the voltage can be assumed to be a constant value V o. Leakage inductance L1 is assumed to be equal to L2. The key waveforms of the BCRR are shown in Fig. 3, and its operation can be described by the following four operating modes [shown in Fig. 4(a)–(d)]. 1) Mode 1 [t0 − t1 ] VGS1 is provided to control Q1 at t0 , and both Q1 and Q2 are turned on during this time interval (Tcharge ). As shown in Fig. 4(a), the voltages across the transformer primaries are thus shorted. Without being forward biased, the four rectifier diodes D1, D2, D3, and D4 are turned off. In addition to being provided by output capacitor

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Fig. 4. Equivalent circuits for the different operation modes of the BCRR.

Co, one-half of the load current is provided by clamping capacitor C1 through C1(+)–S1–L1–R–S2–L2–C1(-). Due to the help of C1, therefore, the ripple of output

Authorized licensed use limited to: National Taiwan Univ of Science and Technology. Downloaded on May 21,2010 at 08:31:02 UTC from IEEE Xplore. Restrictions apply.

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capacitor current Ico is Io/2. Consequently, a smaller output capacitor can be used for the same output-voltage ripple specifications. 2) Mode 2 [t1 − t2 ] As shown in Fig. 4(b), Q2 is turned off at t1 . The voltage across transformer winding P 1 is the sum of the input voltage and the inductor voltage. The input power is transferred to the load via transformer S1 during this interval (Ttransfer ). In addition to providing the load current, part of the input power is used to charge output capacitor Co and clamping capacitor C1 through S1–L1–Co–D2–D1–S1 and S2–D2–D1–C1–L2–S2, respectively. Due to the turning on of D1 and D2, the voltages across D3 and D4 are clamped to VO and VC1 , respectively. 3) Mode 3 [t2 − t1 ] VGS2 is provided to control Q2 at t2 , and both Q1 and Q2 are turned on during this time interval (Tcharge ). Fig. 4(c) shows that the transformer primaries are thus shorted. The four rectifier diodes D1, D2, D3, and D4 are turned off. In addition to being provided by output capacitor Co, one half of the load current is provided by clamping capacitor C1 through C1(+)–S1–L1–R– S2–L2–C1(-). Due to the help of clamping capacitor C1, therefore, the ripple of output capacitor current Ico is Io/2. 4) Mode 4 [t3 − t0 ] As shown in Fig. 4(d), Q1 is turned off at t3 . The voltage across transformer winding P 2 is the sum of the input voltage and the inductor voltage. The input power is transferred to the load via transformer secondary winding S2 during this interval (Ttransfer ). In addition to providing the load current, the input power is used to charge the output capacitor through S2–L2–D4–D3–R– S2 and charge-clamping capacitor C1 through S1–C1– D4–D3–L1–S1, respectively. Due to the turning on of D3 and D4, the voltages across D1 and D2 are clamped to VC1 and V o, respectively. At t0 , Q1 is turned on again to start another switching cycle.

Current waveforms on the transformer secondary winding.

Fig. 6. Normalized rms current of the secondary winding versus the duty cycle of the BCRR and CF-PP-CT.

The turn ratio of the transformer can be calculated as N=

2(1 − Dmax ) · Vo . Vi,min

It can be seen from the previous descriptions that there are two phases, namely, Tcharge and Ttransfer , within each half of the switching cycle. Since duty cycle D is the ON time of Q1 and Q2, the volt–second balance of inductor Lin can be obtained as Vo − V i · TTransfer (1) V i · Tcharge = N Vo 1 V i · (1 − D) · TS . (2) Vi· D− · TS = 2 N Thus, the converter voltage gain can be derived as (3)

where the duty cycle D of each switch must be greater than 50%.

(4)

According to the ampere–second of capacitor C1, Ia is derived as 1 Io · D− (5) · Ts = Ia · (1 − D) · Ts 2 2 Ia =

(2D − 1) Io 4(1 − D)

Io = Io1 + ICo .

III. D ESIGN C ONSIDERATIONS

Vo N = Vi 2(1 − D)

Fig. 5.

(6) (7)

In order to demonstrate the advantage of the full-wave rectifier configuration of the BCRR over that of the CF-PP-CT, the ideal current waveforms of both converters are plotted, as shown in Fig. 5. The value of rms current on one of the transformer second windings can be calculated as IS1,2(rms) (3 − 2 · D) (8) = BCRR Io 8 · (1 − D) IS1,2(rms) 1 . (9) CF-PP-CT = Io 4 · (1 − D) According to (8) and (9), the normalized rms current of one secondary winding versus the duty cycle of the BCRR and CFPP-CT is shown in Fig. 6. As shown, the BCRR has lower secondary winding rms current compared to the CF-PP-CT.



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TABLE I S PECIFICATIONS AND M AIN PARAMETERS OF THE BCRR C ONVERTER

Fig. 8. Oscillograms of the BCRR under low-line and full-load (V i = 16 V and IO = 2 A) operating condition.

Fig. 7. Oscillograms of the BCRR under high-line and light-load (V i = 22 V and IO = 0.2 A) operating condition. Fig. 9. Output voltage and current ripple of the BCRR.

Consequently, the winding conduction loss is reduced in the BCRR. The small-signal model of the proposed converter is similar to that of the current-fed power converter [28]. Consequently, the design of control-related issues, including the compensator design, can be referenced. IV. E XPERIMENTAL R ESULTS A 16–22-V-input, 200-V-output, and 400-W-output converter operating at 150 kHz is implemented. The specifications and key component parameters of the BCRR converter are listed in Table I. Figs. 7 and 8 show the oscillograms of the BCRR operating under high-line–light-load and low-line–full-load conditions. VDS1 and VDS2 are clamped to 2 Vo/N (53 V), as shown in channels 3 and 4 of each diagram. According to the analysis, the rectifier diodes are clamped to the output voltage (200 V) and are shown in channels 5 and 6 of each diagram. Because the leakage energy is absorbed by clamping capacitor C1, all the diodes (D1–D4) are free of voltage spikes.

According to the analysis, the BCRR has the advantage of low output-current ripple over that of the CF-PP-CT. This performance can be verified from the experiment. To operate the full-wave rectifier in the BCRR and CF-PP-CT, the circuitries are implemented by the same hardware with and without the clamping capacitor (C1). The current and output-voltage waveforms of the BCRR and CF-PP-CT are captured under low-line and full-load condition, as shown in Figs. 9 and 10, respectively. Channels 3 and 4 of each diagram show two parts of the output current, namely, the current provided by the full-wave rectifier circuit (Io1 ) and the current provided by the output capacitor (Ico ). As shown in channel 5 of each diagram, less than 1-V output-voltage ripple is obtained in both circuits. Due to the help of the clamping capacitor, however, one 68-μF/450-V capacitor is used in the BCRR instead of two 68-μF/450-V capacitors in the CF-PP-CT. The number of electrolytic capacitors can be minimized, resulting in increased power density.


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Fig. 10. Output voltage and current ripple of the CF-PP-CT.

Fig. 12.

Efficiency measurement of the power stage of the BCRR.

with a small turn ratio (7.5 = 15/2) transformer, and problems caused by these parasitic components are alleviated. In addition, two series-connected voltage-clamped diodes have been used to replace one high-voltage-rating diode. Also, the secondary leakage energy has been recycled so that the secondary rectifier diodes are free of voltage spikes, and each diode is thus clamped to V o. Consequently, a low-voltage-rating rectifier diode can be used to reduce the conduction loss. Due to the help of the clamping capacitor, moreover, the output filter current ripple is reduced, resulting in the use of a smaller number of highoutput-voltage-rating electrolytic capacitors for the same output power. Finally, the rms current on the secondary winding is lower, and the conduction losses can be reduced. These advantages make the proposed converter suitable for FC-powered applications. Fig. 11. Efficiency comparisons between the BCRR and CF-PP-CT.

Fig. 11 shows efficiency comparisons between the BCRR and CF-PP-CT operated at low-line conditions. The CF-PP-CT has higher efficiency because the switching loss is dominant to the total losses at light-load operating conditions. On the contrary, the BCRR has higher efficiency at heavy-load operating conditions due to the reductions of conduction and snubber losses. Fig. 12 shows the measured efficiency of the proposed converter with different input-voltage and load conditions. Unlike other pulse width modulation converters, the proposed BCRR has lower efficiency at low-line conditions because the conduction losses are dominantly caused by high input currents. As shown, a maximum efficiency of 93.2% occurs at 22-V-input and 200-V/1.6-A-output operating condition. V. C ONCLUSION The analysis, design, and experiment of the proposed BCRR circuit have been presented. Due to its voltage-boost property, the high voltage ratio (12.5 = 200 V/16 V) can be obtained

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Ching-Shan Leu (M’96) received the B.S. degree from the Department of Electrical Engineering, National Taiwan Oceanic University, Keelung, Taiwan, in 1974, the M.S. degree from the Department of Electrical and Computer Engineering, The Ohio State University, Columbus, in 1985, and the Ph.D. degree from the Department of Electrical and Computer Engineering, Virginia Polytechnic Institute and State University, Blacksburg, in 2006. From 1978 to 2003, he was with Chun-Shan Institute of Science and Technology, Longtan, Taiwan, as an Assistant Researcher. Since 2006, he has been with the Department of Electrical Engineering, National Taiwan University of Science and Technology, Taipei, Taiwan, where he is currently an Assistant Professor. His research is focused on high-frequency power converter technology for renewable energy applications.

Ming-Hui Li was born in Taoyuan, Taiwan, on December 21, 1981. He received the B.S. degree from the Department of Electrical Engineering, National United University, Miaoli, Taiwan, in 2003, and the M.S. degree from the Department of Electrical Engineering, National Taiwan University of Science and Technology, Taipei, Taiwan, in 2009. His research interests include switching-mode dc–dc power converters. He is currently with Skynet Electronics Company, Ltd., Taipei.
