A Novel Dual Full-Bridge LLC Resonant Converter for

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Jun 26, 2017 - charge, hence no reverse recovery problems; and. 108. 5) High ...... voltage is always constant and there is no tolerance of resonant 531.
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS

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A Novel Dual Full-Bridge LLC Resonant Converter for CC and CV Charges of Batteries for Electric Vehicles (July 2017) Hai-Nam Vu and Woojin Choi, Member, IEEE

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Index Terms—CC/CV charge, dual full-bridge LLC (FBLLC) resonant converter, high efficiency, variable resonant network.

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I. INTRODUCTION

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LECTRIC vehicles (EVs) and Plug-in hybrid electric vehicles (PHEVs) are becoming more popular in an effort to guarantee energy security and to lower fuel consumption and emissions. On-board chargers (OBCs) with high power density and high power conversion efficiency are crucial for the proliferation of EVs and PHEVs. An OBC is normally composed of an ac–dc converter followed by a dc–dc converter to charge batteries from a dc link. One of the most popular dc–dc converters for battery chargers is the full-bridge LLC (FBLLC) resonant converter. However, the FBLLC resonant converter needs to utilize one of the three control methods, such as frequency

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Q3

E

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modulation, phase-shift modulation, or a modified structure in order to implement CC and CV charges for batteries. In [1]–[13], frequency modulation methods are used to control the output voltage or output current of an FBLLC resonant converter. The LLC resonant converter with frequency modulation has been researched in many aspects. The research works in [1]–[5] focus on the design of the LLC resonant converter for a wide output voltage range such as the battery charge applications. In [6], it is suggested to use the LLC resonant converter for the distributed power system operating at a high switching frequency due to its ZVS turn-on characteristics. In [7], a design methodology for the LLC resonant converters by using a capacitor-diode clamp is proposed to limit the current in the overload conditions. For the high-power applications, an interleaved LLC resonant converter is suggested to reduce the output voltage ripple [8] and to limit the effects of the component tolerances [9]. Some methods to track the maximum efficiency point of the LLC resonant converter based on the state-of-charge of a battery are introduced in [10] and [11]. In [12] and [13], optimal design of LLC resonant converter is discussed in relation to the dead time, the maximum switching frequency, and the time-weighted average efficiency. However, all of the methods mentioned in [1]–[13] are not able to implement the CC/CV charge of the battery without the wide range of the switching frequency variation. Once the resonant converter operates out of the resonant frequency, it loses its main advantages such as a low switching loss and a low circulating current. As a result, it becomes less efficient. When the switching frequency is higher than the resonant frequency, the primary switches of the FBLLC resonant converter suffer from a high turn-off current. When the switching frequency is lower than the resonant frequency, the circulating current occurs in the primary side. In addition, the wide range of the switching frequency variation causes difficulty in optimizing the magnetic components [14]. Besides, higher conduction losses and a lower power density of the converter are unavoidable when it is not optimized. In [15]–[19], a phase-shift modulation method is used to increase the efficiency of an FBLLC resonant converter under a light load condition. Compared to the frequency modulation method, phase-shift modulation can reduce the peak-to-peak flux density of the transformer. As a result, the core loss of the transformer can be significantly reduced and a high conversion efficiency can be achieved at a light load. However, the

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Abstract—This paper introduces a novel concept for implementing CC and CV charges of batteries for electric vehicles using a dual full-bridge LLC (FBLLC) resonant converter, which shares its primary switches. In the proposed concept, the CC and CV charges are implemented using the inherent characteristics of an LLC converter as a current source and a voltage source. One FBLLC resonant converter operates with a fixed-resonant network whereas the other operates with a variable resonant network for the CC and CV charge operations. The proposed converter can achieve ZVS and nearly ZCS for all of the primary switches in the CC charge operation, and ZVS for all of the primary switches in the CV charge operation, which lead to a high efficiency. In addition, fixed switching frequency operation is possible in both the CC and CV charges thanks to the variable resonant network. A 3.3-kW prototype converter is implemented to verify the feasibility and validity of the proposed converter and a maximum efficiency of 98% was achieved.

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Manuscript received February 15, 2017; revised May 1, 2017 and June 26, 2017; accepted July 12, 2017. (Corresponding author: Woojin Choi.) H.-N. Vu was with the Department of Electrical Engineering, Soongsil University, Seoul 156-743, South Korea. He is now with the Centre for Technology Innovation, Hanoi University of Science and Technology, Hanoi, Vietnam (e-mail: [email protected]). W. Choi is with the Department of Electrical Engineering, Soongsil University, Seoul 156-743, South Korea (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIE.2017.2739705

0278-0046 © 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications standards/publications/rights/index.html for more information.

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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS

Fig. 2.

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Vo (s) = n

1+

Vac

L r s+ C 1r s Lm s

 + Rac Lr s +

1 Cr s



fr 2 = Io (s) =







n Lr s +

1 Cr s

 1 + Rac

132 133 134 135 136 137 138 139

140 141 142 143 144 145 146 147 148 149

(3)

(Lm + Lr )Cr .

130 131

(2)

1 Vin

129

.

The lower resonant frequency fr 2 is determined by the resonant components Lr and Cr plus the magnetizing inductance Lm , as shown in (3). When the FBLLC resonant converter operates at fr 2 , its output current is independent of the load variation since the term associated with the load resistance Rac becomes zero in the denominator as shown in (4). The FBLLC resonant converter works as a constant current source at fr 2 . However, the lower resonant frequency is rarely used since the converter can achieve ZCS turn-off but loses ZVS turn-on, which is not preferred for power MOSFETs [22].

II. PROPOSED DUAL FBLLC RESONANT CONVERTER

It is well known that the LLC resonant converter can operate as a voltage source or as a current source at certain resonant frequencies. The proposed dual FBLLC resonant converter (see Fig. 1) utilizes these inherent characteristics to implement the CC and CV charges. The CC charge can be implemented by operating the upper and lower FBLLC converters as a voltage and current source, respectively. The CV charge can be implemented by operating both of the converters as voltage sources. The basic characteristics of the FBLLC converter are summarized in the Section II-A. The structure and operating principle of the proposed converter are described in Section II-B. Sections II-C and II-D show the methods for implementing the CC and CV charge operations and designing the resonant tanks of the proposed converter, respectively.

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Fig. 2 shows an equivalent circuit of a single FBLLC resonant converter that was derived by first harmonic approximation, and it has two resonant frequencies [2], [3]. The higher resonant frequency fr 1 is determined by the resonant components Lr and Cr , as in (1). When the FBLLC resonant converter operates at fr 1 , its voltage gain is independent of the load variation since the term associated with the load resistance Rac becomes zero in the denominator as shown in (2) [2]. Hence, the FBLLC resonant converter works as a constant voltage source at fr 1 , and the primary switches can achieve the ZVS turn-on 1 √ fr 1 = (1) 2π Lr Cr

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A. Two Resonant Frequencies of a Single FBLLC Resonant Converter

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phase-shift modulation method is not preferred for heavy loads due to its high turn-off currents for the primary switches when compared with the frequency modulation method. In order to avoid the above-mentioned problems, modified structures of an FBLLC resonant converter are presented in [20]–[21]. These FBLLC resonant converters can transform their structures from full-bridge to half-bridge and vice versa by using additional switches. Hence, it is possible to regulate the output voltage with a minimal operating frequency variation and to optimize the design of the magnetic components. However, these kinds of converters require two additional active switches, which operate with hard switching, to transform their structure and all of the active switches suffer from high turn-off currents leading to a lower efficiency. In this paper, a novel dual FBLLC resonant converter is proposed to cope with all the aforementioned problems as shown in Fig. 1. The advantages of the proposed converter can be summarized as follows: 1) ZVS and nearly ZCS for all primary switches in CC charge, and ZVS for all primary switches in CV charge; 2) Only one additional switch to transform the resonant tank and no switching loss associated with it; 3) A fixed-frequency resonant operation in both CC and CV charge and hence lower circulating energy; 4) Perfect soft switching of all the rectifier diodes due to the resonant operation over the entire CC and CV mode charge, hence no reverse recovery problems; and 5) High voltage gain and lower voltage rating of the rectifier diodes due to the series connection of two transformers and two rectifier bridges.

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Proposed dual full-bridge LLC resonant converter.

of

Fig. 1.

AC equivalent circuit of a single FBLLC converter.

1 Lm s L r s+ C 1r s

 +1 .

(4)

B. Structure and Operating Principle of the Proposed Converter The above-mentioned characteristics of the FBLLC converter can be utilized to implement the CC and CV charges of the OBC of EVs. The proposed converter is composed of two FBLLC converters, which share primary switches as shown in Fig. 1. In the proposed converter, the primary switches always operate with a fixed frequency and a fixed duty cyle of 50%. Whereas the FBLLC1 converter is designed to always work as a voltage source at the higher resonant frequency, and the FBLLC2 converter has a variable structure with an additional switch to

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VU AND CHOI: NOVEL DUAL FB LLC RESONANT CONVERTER FOR CC AND CV CHARGES OF BATTERIES FOR ELECTRIC VEHICLES (JULY 2017)

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TABLE I RESONANT FREQUENCIES USED TO IMPLEMENT THE CC AND CV CHARGE OF THE PROPOSED CONVERTER CC mode charge

CV mode charge

Used

Used

Not Used

Not Used

Not Used

Not Used

fr 2 L L C 2 = 1 √

Used

Not Used

L lk 2 , L m 2 , fr 1 L L C 3 = 1 √ (C r 2 + C r 3 )

Not Used

Used

Not Used

Not Used

L lk 1 , L m 1 , fr 1 Cr 1

LLC1

=





1 L lk 1 C r 1

fr 2 L L C 1 = 1 √ 2π

FBLLC2 converter

L lk 2 , L m 2 , fr 1 Cr 2





(L m 1 + L l k 1 )C r 1

LLC2

=

L l k 2 (C r 2 + C r 3 )

164 165 166 167 168 169 170 171 172 173 174 175 176 177 178 179 180 181 182 183 184

185 186 187 188 189 190 191 192 193

1

(L m 2 + L l k 2 )(C r 2 + C r 3 )

Fig. 3. AC equivalent circuit of the proposed dual FBLLC resonant converter in the CC mode operation.

transform its resonant tank, and hence the resonant frequency, to make it work as a current source for CC charge and as a voltage source for CV charge. Table I shows three different resonant tanks formed in the proposed dual FBLLC resonant converter, their resonant frequencies, and their usages in implementing the CC and CV charge. The upper converter has a resonant tank composed of Llk1 , Lm 1 , and Cr 1 . Only the resonant operation at the higher resonant frequency fr 1 LLC1 is used for both the CC and CV charges. The lower converter has two resonant tanks, one composed of Llk2 , Lm 2 , and Cr 2 , and the other composed of Llk2 , Lm 2 , and (Cr 2 + Cr 3 ). In the CC charge, the lower converter operates at the lower resonant frequency fr 2 LLC2 so that it can work as a current source. In the CV charge, it operates at the higher resonant frequency fr 1 LLC3 so that it can work as a voltage source by turning ON an additional switch S M o de . Here, the resonant network of the FBLLC2 converter is designed to have a lower resonant frequency fr 2 LLC2 , which is the same as the higher resonant frequency fr 1 LLC1 of the upper resonant converter. In the CV charge, both the upper and lower converters work as a voltage source by the resonant operation at fr 1 LLC1 and fr 1 LLC3 , respectively. In this mode, an additional switch is turned ON to transform the resonant tank of the FBLLC2 converter to resonate at fr 1 LLC3 .

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L lk 2 C r 2

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(L m 2 + L l k 2 )C r 2

fr 2 L L C 3 = √ 2π



1

of

FB LLC1 converter

Higher and lower resonant frequencies of each resonant tank

ro

Resonant tank

C. Design Considerations for the CC Mode Operation of the Proposed Converter As previously mentioned, the CC charge can be implemented by operating the FBLLC1 resonant converter as a voltage source and by operating the FBLLC2 resonant converter as a current source. Hence, the FBLLC1 resonant converter and the FBLLC2 resonant converter need to resonate at fr 1 LLC1 and fr 2 LLC2 , respectively. It should be noted that the FBLLC1 converter must be designed to have a lower output voltage than that of the

Fig. 4. Voltage gain of FBLLC1 and output current of FBLLC2 curves w.r.t. the switching frequency variation under different load conditions.

minimum battery voltage (250 V in this research) in order to operate the entire converter as a current source. The switch S M o de is turned OFF during the CC charge operation. In this case, the resonant tank of the FBLLC2 resonant converter is composed of Llk2 , Lm 2 , and Cr 2 . The equivalent circuit of the proposed converter in the CC charge can be drawn as shown in Fig. 3. The higher resonant frequency of the FBLLC1 resonant converter fr 1 LLC1 and the lower resonant frequency of the FBLLC1 resonant converter fr 2 LLC2 can be calculated, respectively, by the following equations: fr 1

LLC1

=

fr 2

LLC2

=

1 2π Llk1 Cr 1 2π



(5)

1  . (Lm 2 + Llk2 )Cr 2

(6)

Since the two converters share the primary switches, the resonant tank of two converters need to be designed to have the same resonant frequency, as shown in the following equation: fr 1

LLC1

= fr 2

LLC2 .

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205 206 207

(7)

Fig. 4 shows the voltage gain curves for the FBLLC1 resonant converter and the output current curves for the FBLLC2 resonant

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converter under different load conditions. As shown in Fig. 4, the FBLLC1 converter and the FBLLC2 converter both resonate at a switching frequency fs , thereby operating the FBLLC1 resonant converter as a constant voltage source and the FBLLC2 resonant converter as a constant current source, respectively. As shown in Fig. 3, the output of the FBLLC1 resonant converter is represented by a constant voltage source Vo1 , which can be calculated by

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Vo1 = n1 Vin 219 220 221

where n1 is the turns ratio of the transformer TR1 of the FBLLC1 resonant converter. The output of the FBLLC2 resonant converter can be represented by a constant current source Io2 , which is calculated by Io2 =

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Vin

  n2 ωs Llk2 −

1 ωs Cr 2

  .

Io = Io2 .

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(10)

Therefore, the output power of the FBLLC1 resonant converter and FBLLC2 resonant converter can be represented by PFBLLC1 = Vo1 Io , PFBLLC2 = (Vo − Vo1 )Io .

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(9)

In the CC charge, the output of the proposed converter is composed of a voltage source and a current source connected in series. The output current of the FBLLC2 converter is the same as the charge current of the battery in the CC charge as shown in the following equation:

IEE

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(8)

Fig. 6. Voltage gain curves of the FBLLC1 and the FBLLC2 converters w.r.t. the switching frequency variation under different load conditions.

FBLLC2 resonant converter can be expressed by fr 1

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218

of

Fig. 5. AC equivalent circuit of proposed converter in CV charge operation.

(11)

D. Design Considerations for the CV Mode Operation of the Proposed Converter In order to implement the CV charge of the proposed converter, both converters work as voltage sources. Since the resonant tank of the FBLLC2 used for the CC charge is designed to satisfy (7), the higher resonant frequency of it fr 1 LLC2 = √ 1 cannot be the same as the higher resonant frequency 2π L l k 2 C r 2 of the FBLLC1 resonant converter fr 1 LLC1 . Hence, the switch S M o de is turned ON to transform the resonant tank in this mode. Since the resonant capacitor Cr 2 and Cr 3 are connected in parallel, the total capacitance increases to (Cr 2 + Cr 3 ) as shown in Fig. 5. Hence, in this mode, the resonant frequency of the

LLC3

=

1  . 2π Llk2 (Cr 2 + Cr 3 )

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(12) 242

Since both the converters have to resonate at the same switching frequency in the CV charge operation, the higher resonant frequency of the FBLLC2 resonant converter fr 1 LLC3 needs to be designed to have the same resonant frequency as the higher resonant frequency of the FBLLC1 resonant converter fr 1 LLC1 as shown in the following equation: fr 1

LLC3

= fr 1

LLC1 .

246 247 248

249 250 251 252 253 254 255

(14)

where n2 is the turns ratio of the transformer TR2 of the FBLLC2 resonant converter. In this CV charge, the output of the proposed converter is composed of two voltage sources connected in series and the output voltage can be represented by Vo = Vo1 + Vo2 = Vin (n1 + n2 ) .

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(13)

Fig. 6 shows the voltage gain curves for the FBLLC1 resonant converter and the FBLLC2 resonant converter under different load conditions. As shown in Fig. 6, the FBLLC1 converter and the FBLLC2 converter both resonate at a switching frequency fs , thereby operating both of them as constant voltage sources. The output of the FBLLC2 resonant converter can be represented by a constant voltage source Vo2 which is calculated by Vo2 = n2 Vin .

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(15)

E. Charge Sequence of the Proposed Converter

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The specification of the proposed converter is shown in Table II, and the CC/CV charge profile of the EV battery and its equivalent impedance during the charge are shown in Fig. 7. The battery pack is composed of 100 Li-ion battery cells and

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VU AND CHOI: NOVEL DUAL FB LLC RESONANT CONVERTER FOR CC AND CV CHARGES OF BATTERIES FOR ELECTRIC VEHICLES (JULY 2017)

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TABLE II RESONANT SPECIFICATION OF THE PROPOSED CONVERTER Parameter Input voltage Output voltage (battery voltage) Power rating CC charge current CV charge voltage Resonant frequency

Designator

Value

V in Vo Po Io Vo fr

380–420 V 250–420 V 3.3 kW 7.8 A 420 V 50 kHz

of

Fig. 8. Power distribution FBLLC1 and FBLLC2 resonant converter in CC and CV mode operation.

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III. MODES OF OPERATION OF THE PROPOSED DUAL FBLLC RESONANT CONVERTER

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In this section, the modes of operation of the proposed converter in the CC and CV charges are detailed. Some of the circuit parameters are predefined as follows. The angular switching frequency and the switching period are expressed by (16) and (17), respectively

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the voltage of it varies from 250 to 420 V. The charge starts with CC mode charge with a constant current of 7.8 A. The CC mode charge is finished when the voltage of the battery pack reaches its maximum voltage (420 V). After then, the proposed converter switches its charge mode to CV mode charge by turning ON the switch S M o de . The CV mode is ended when the charge current is decreased to 0.78 A (0.1 C). Here, the output voltage of the FBLLC1 converter is designed to take a half of the maximum battery voltage (210 V), which is less than the minimum battery voltage (250 V). Therefore, in the CC mode the charge current depends solely on the output current of the FBLLC2 resonant converter. The FBLLC1 converter must be designed to have a lower output voltage than that of the minimum battery voltage (250 V) in order to operate the entire converter as a current source. Otherwise constant current mode cannot be implemented since the FBLLC1 becomes active due to its higher voltage than that of the battery.

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Fig. 7. CC/CV charge profile of the EV battery pack and its equivalent impedance during the charge.

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voltages have the same values. Since the maximum output powers of both resonant converters are the same, the components with same ratings can be used for each converter, which is advantageous.

F. Power Distribution in FBLLC1 Resonant Converter and FBLLC2 Resonant Converter Since the output of two FBLLC resonant converters is connected in series, the output power of each FBLLC resonant converter is proportional to its output voltages in both CC and CV mode operations. The power distribution in two converters during CC and CV mode operations is shown in Fig. 8. During the CC mode charge operation, the output power of FBLLC1 is 1638[W] (50% of the total output power) and the output power of FBLLC2 increases gradually from 312[W] (19% of the total output power) to 1638[W] (50% of the total output power). In the CV mode operation, the output powers of FBLLC1 and FBLLC2 are equal when their output

ωs = 2πfs Ts =

1 . fs

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(16) (17)

A. CC Charge Operation

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The switch S M o de is always turned OFF during the CC charge operation. There are four operating modes in a half switching cycle of the proposed converter in the CC charge operation. The key waveforms for each of the operating modes are illustrated in Fig. 9. Mode 1 (t0 –t1 ): Switches S2 and S3 are turned OFF with nearly ZCS at t0 . The primary current of the FBLLC resonant converter ipri discharges the output capacitance of the switches S1 and S4 . After the voltage of the switches S1 and S4 drops to zero, the body diodes of these switches conduct and achieves zero-voltage switching conditions. The slope of the magnetizing current im 1 (t) can be expressed by

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Vin dim 1 (t) = . dt Lm 1

4Vin sin (ωS (t − t1 ) − θpri1 ) π |Zin1 (jωS )|

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(18)

Mode 2 (t1 –t2 ): The switches S1 and S4 are turned ON with ZVS at t1 . The resonant tank of the FBLLC1 resonant converter, which is composed of Llk1 and Cr 1 , starts to resonate and the primary current ipri1 (t) can be expressed by ipri1 (t) =

308

(19)

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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS

Therefore, the summation of the above two primary currents expressed by (19) and (20) is the same as the primary switch current ids1 as shown in Fig. 7. Mode 3 (t2 –t3 ): At t2 , the magnitude of the primary current ipri2 (t2 ) is equal to that of the magnetizing current of the transformer TR2 in the FBLLC2 resonant converter. The power transfer through it ends and the rectifier diode D5 achieves ZCS turn-off. In this mode, the primary currents ipri1 and ipri2 can be expressed by (19) and (20), respectively. The magnetizing current im 2 (t) can be expressed by Vo2 (t − t2 ) . n2 Lm 2

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N =1,3,5...

4Vin sin (N ωS (t − t1 ) N π |Zin2 (jN ωS )|

− θpri2 N ) 333 334 335 336 337

(20)

where Zin2 (jN ωS ) is the input impedance of the FBLLC2 resonant converter at the angular switching frequency N ωS , and θpri2 N is the phase angle of the N th component of the primary current ipri2 with respect to that of the N th component of the voltage vab .

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B. CV Charge Operation

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The switch S M o de is turned ON during the CV charge operation. In this operation, both the FBLLC1 and FBLLC2 converters work at their higher resonant frequencies. The operating principle in CV charge operation is the same as the conventional FBLLC resonant converter and it can be found in [2]. In this mode, if the converter operates at the frequency higher than 31 kHz, the converter operates in inductive region and the ZVS condition for all of the primary switches can be guaranteed regardless of the load variation.

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IV. DESIGN PROCEDURE FOR RESONANT NETWORKS

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In this section the design procedure for the resonant tanks is detailed. In order to implement the CC and CV charges with fixed-frequency operation, the resonant tanks need to be designed carefully by the following procedure. Step 1 (Design of magnetizing inductance Lm 2 ): In CC charge operation, the FBLLC2 resonant converter operates at its lower resonant frequency fr 2 LLC2 . Therefore, as shown in Fig. 3 and (6), the impedance of the resonant tank in FBLLC2 becomes zero at the angular switching frequency ωs and (22) can be derived    1   . (22) |ωs Lm 2 | = ωs Llk2 − ωs Cr 2 

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ipri2 (t) =

∞ 

342

348

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where Zin1 (jωS ) is the input impedance of the FBLLC1 resonant converter at the angular switching frequency ωS and θpri1 is the phase angle of the primary current ipri1 with respect to that of the fundamental component of the voltage vab . At the same time, the resonant tank of the FBLLC2 resonant converter, which is composed of Lm 2 , Llk2 , and Cr 2 , starts to resonate and the primary current ipri2 can be expressed by (20). Since the FBLLC2 resonant converter operates at the lower resonant frequency fr 2 LLC2 , its primary current can be expressed by the Fourier series in (19)

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Mode 4 (t3 –t4 ): At t3 , the switches S1 and S4 are turned with nearly ZCS, and the rectifier diode D1 achieves ZCS turn-off. As mentioned in Mode 2, since the switch current is a summation of the two primary currents of the FBLLC resonant converters, some design rules should be applied to design the transformer TR1 of the FBLLC1 resonant converter to achieve nearly ZCS turn-off of the primary switches. This is described in detail in the following section. Since the operation of the proposed converter during the second half of the switching cycle is symmetric to that during the first half of the switching cycle, it is omitted here. OFF

Fig. 9. Key waveforms of the proposed converter in CC charge operation.

339 340

(21)

of

im 2 (t) = im 2 (t2 ) −

338

By using (9), (10), (14), and (22), the magnetizing inductance Lm 2 of FBLLC2 converter can be calculated by Lm 2 =

Vin . n2 ωs Io

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(23)

Step 2 (Design of leakage inductance Llk2 ): In the FBLLC2 resonant converter, the rms value of the primary current ipri2

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VU AND CHOI: NOVEL DUAL FB LLC RESONANT CONVERTER FOR CC AND CV CHARGES OF BATTERIES FOR ELECTRIC VEHICLES (JULY 2017)

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the value of Llk2 . Therefore, the magnetizing inductance Llk2 should be designed to be as large as possible in order to minimize the rms values of the primary current ipri2 RM S , which results in lower conduction losses of the primary switches. A leakage inductance value larger than 120 μH is suitable to keep the rms value of the primary current ipri2 less than 5.5 A. Step 3 (Design of resonant capacitor Cr 2 ): The resonant capacitor Cr 2 can be calculated from (6) as Cr 2 =

1 ωs2

(Lm 2 + Llk2 )

.

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It can be seen from (24) that the magnitude of Ipri2 RM S depends on the values of |Zin2 (jN ωS )|. The input impedance of the FBLLC2 resonant converter at the angular switching frequency N ωS can be expressed by

of

Zin2 (jN ωs ) =

1 + jN ωs Llk2 jN ωs Cr 2

+

392 393 394

n22 π82

jN ωs Lm 2 Rac2 jN ωs Lm 2 + Rac2

(25)

o2 Ro V o 1V+V o2

where Rac2 = . Since Cr 2 can be calculated by 1/(ωs 2 (Lm 2 + Llk2 )) at the lower resonant frequency, the input impedance Zin2 (jN ωS ) can be expressed by

IEE

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Zin2 (jN ωs ) =

1 jN ωs ω 2 (L m 21+L l k 2 ) s

+ jN ωs Llk2 +

395 396 397 398 399 400 401 402 403 404

− Cr 2 .

jN ωs Lm 2 Rac2 . (26) jN ωs Lm 2 + Rac2

Equation (26) shows that the input impedance Zin2 (jN ωs ) depends on Llk2 and Lm 2 once the load impedance is designated. Since Lm 2 is already decided in step 1, the rms value of the primary current Ipri2 RM S only depends on the value of Llk2 . Fig. 10 illustrates the relationship between the leakage inductance Llk2 and the rms current value of the primary current ipri2 of the FBLLC2 resonant converter under a certain operating condition (VDC = 400 V, Vo = 250 V, Po = 1.95 kW, Io = 7.8 A, and fs = 50 kHz). As shown in Fig. 10, the rms values of the primary current ipri2 is inversely proportional to

Cr 1 =

1 ωs2 Llk1

.

411 412

413 414

415 416 417 418 419 420 421 422 423 424 425

426 427 428 429 430 431 432 433 434 435 436 437 438 439 440

(30)

The total inductive energy EL stored in the primary side at the switching moment can be calculated by EL =

409 410

(29)

Step 7 (Design of magnetizing inductance Lm 1 ): In the CC charge operation of the proposed converter, the FBLLC1 and FBLLC2 converters operate in the ZVS and ZCS regions, respectively. In order to ensure the soft switching at both the turn-on and turn-off of the switches, the proposed converter, which is a combination of two FBLLC converters, should operate in the ZVS region over the entire load range during the CC charge. In this section, the magnetizing inductance Lm 1 is designed to ensure ZVS and nearly ZCS for all of the primary switches over the entire load of the CC charge. The ZVS for all of the primary switches can be achieved by designing the inductive energy EL stored in the primary side to have a higher value than the capacitive energy EC stored in the output capacitances of the primary switches by the following equation: EL ≥ E C .

408

(28)

ro

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depends on the leakage inductance Llk2 . The rms value of the primary current ipri2 can be calculated from (20) by the following equation:    ∞  4Vin √ . (24) Ipri2 RM S = 2N π |Zin2 (jN ωS )| N =1,3,5...

1

ωs2 Llk2

Step 5 (Design of leakage inductance Llk1 ): There are no specific requirements for the design of the leakage inductance Llk1 since the FBLLC1 converter always operates at its higher resonant frequency and only the magnetizing current contributes to the discharge of the output capacitors of the primary switches. However, it is better to minimize this since the voltage stress applied to the resonant capacitor Cr 1 is increased when the value of the leakage inductance Llk1 is large. Step 6 (Design of resonant capacitor Cr 1 ): The resonant capacitor Cr 1 can be calculated using (29) after determining the leakage inductance Llk1 in the previous step

EP

384

Cr 3 =

RMS value of the primary current ip ri2 w.r.t. L lk 2 value.

406 407

(27)

Step 4 (Design of resonant capacitor Cr 3 ): The resonant capacitor Cr 3 can be calculated from (12) as Fig. 10.

405

441 442

1 1 1 Lm 1 i2m 1 (t3 ) + Llk1 i2pri1 (t3 ) − Lm 2 i2m 2 (t3 ) 2 2 2 1 − Llk2 i2pri2 (t3 ) . (31) 2

where im 1 (t3 ) = 4LVmi n1 T s , ipri1 (t3 ) = im 1 (t3 ), ipri2 (t3 ) =

∞ 4V i n Vo 2 Ts N =1,3,5... N π |Z i n 2 (j N ω S )| sin (θpri2 N ), im 2 (t3 ) = n 2 L m 2 4

443 444

8

446 447 448

449 450

− n 2VLo m2 2 T2s ( π −πθ m 2 ), and θm 2 is the phase angle of the primary voltage vpri2 with respect to that of the voltage vab . The capacitive energy stored in the output capacitors of one pair of primary switches EC can be calculated by 1 EC = 2 × Coss Vin2 . (32) 2 By using (30)–(32), the magnetizing inductance Lm 1 can be calculated by 

4Ts 2 2 2 2 2Coss Vin + Lm 2 im 2 (t3 ) + Llk2 i (t3 ) Lm 1 Vin − Lm 1 − Llk1 ≤ 0.

452 453 454 455 456 457 458 459 460 461 462

Since all of the resonant network parameters for the FBLLC2 and FBLLC1 converters have been decided through step 1 to step 6, the inductive energy EL only depends on the magnetizing inductance Lm 1 . In order to make the turn-off currents of the primary switches as small as possible, the magnetizing inductance Lm 1 needs to be as large as possible in order to achieve a nearly ZCS condition. Therefore, the magnetizing inductance Lm 1 can be calculated from (33) as (34) as shown at the bottom of this page. In order to verify the full soft-switching condition during the CC mode operation, the input impedance Zin (s) of the proposed converter needs to be drawn by using

463 464 465 466

Zin1 (s) Zin2 (s) . Zin1 (s) + Zin2 (s)

The input impedance of the FBLLC1 resonant converter Zin1 (s) and the input impedance of the FBLLC2 resonant converter Zin2 (s) can be calculated by using (36) and (37), respectively Zin1 (s) = sLlk1 +

468 469

470 471 472 473 474

1 sLm 1 Rac1 + sCr 1 sLm 1 + Rac1

1 sLm 2 Rac2 + + sCr 2 sLm 2 + Rac2

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Zin2 (s) = sLlk2 467

(35)

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Zin (s) =

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451

(33)

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(36) (37)

where Rac1 and Rac2 are the reflected load resistance of the FBLLC1 converter and FBLLC2 converter, respectively, and it can be expressed by (38) and (39), respectively Rac1 = n21

8 Vo1 Ro 2 π Vo1 + Vo2

(38)

Rac2 = n22

8 Vo2 Ro . 2 π Vo1 + Vo2

(39)

Fig. 11 shows the input impedance Zin of the proposed converter with varying battery impedance during the CC mode operation drawn with the specification in Table II and the circuit parameters in Table II. As shown in Fig. 11, at the switching frequency, the phase of input impedance Zin always lies in

Fig. 11. Input impedance variation of the proposed converter with varying battery impedance during the CC mode operation. (a) Phase of input impedance. (b) Magnitude of input impedance.

between 3°–8° over the wide range of battery impedance variation. Therefore, the proposed converter can achieve ZVS and nearly ZCS for all of the primary switches during the CC mode operation. V. IMPACT OF THE TOLERANCES IN RESONANT COMPONENTS AND CLOSED-LOOP CONTROL SCHEME OF THE PROPOSED CONVERTER

Lm 1 =

476 477 478

479 480 481

A. Impact of the Tolerances in Resonant Components

482

One important factor needed to be considered in implementing the proposed converter is the tolerances of resonant components, which can affect the output current gain and output voltage gain of the converter. In the design procedure shown in Section IV, the leakage inductance and the magnetizing inductance of the transformer are measured after the transformer is actually implemented. Then, the values of the resonant capacitors (Cr 1 , Cr 2 , and Cr 3 ) are calculated again depending

483

 1+

475

   2 s 1 + Llk1 2Coss Vin2 + Lm 2 i2m 2 (t3 ) + Llk2 i2pri2 (t3 ) 8T Vin    2 s 2 2Coss Vin2 + Lm 2 i2m 2 (t3 ) + Llk2 i2pri2 (t3 ) 4T . Vin

(34)

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VU AND CHOI: NOVEL DUAL FB LLC RESONANT CONVERTER FOR CC AND CV CHARGES OF BATTERIES FOR ELECTRIC VEHICLES (JULY 2017)

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TABLE III VARIATION IN THE RESONANT FREQUENCIES DUE TO THE TOLERANCE OF THE RESONANT CAPACITORS

ro

Fig. 13. Variation in the output voltage gain of the proposed converter according to the variation in the resonant frequencies.

Fig. 12. Change of output current according to the variation of resonant frequencies.

494 495 496 497 498 499 500 501 502 503 504 505 506 507 508 509 510 511 512 513 514 515 516 517 518 519 520 521

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on the leakage inductances and the magnetizing inductances of the transformers, and the actual resonant capacitors are selected by using the inductance values and the switching frequency as shown in (27)–(29). Therefore, the resonant frequencies fr 1 L L C 1 , fr 2 L L C 2 , and fr 1 L L C 3 are only affected by the tolerance of the resonant capacitors. The following analysis shows as to how much the tolerance of the resonant capacitors affects the variation in the output current and output voltage gains. If the resonant capacitors have ±5% tolerance in theirs values (typical value of J-type), the resulting variation in the resonant frequencies can be calculated as shown in Table III. When the value of resonant frequency varies from 97.6% (48.8 kHz) to 102.6% (51.3 kHz), the output current gain in CC mode operation is varied from 95% (7.4 A) to 105% (8.2 A), as shown in Fig. 12. Similarly, when the value of resonant frequency varies from 97.6% (48.8 kHz) to 102.6% (51.3 kHz), the output voltage gain in CV mode operation is varied from 102% (428.4 V) to 98% (411.6 V), as shown in Fig. 13. As shown in Figs. 12 and 13, the proposed converter still exhibits desired characteristics even though the resonant capacitors have a small deviation in their values. In the CC mode operation, it is important to keep the current constant and the switching frequency needs to be adjusted to find out the exact resonant frequency for the FBLLC2 converter. In this case, the output voltage of the FBLLC1 may not be exactly 210 V but it is not important as long as the output voltage of FBLLC1 is lower than that of the battery. In the CV mode operation, switching frequency needs to be adjusted to keep the output voltage (420 V) constant and the output voltage of each converter may not be exactly 210 V. In this case, a closed-loop control is essential to

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491 492

Fig. 14.

Closed-loop control scheme of the proposed converter.

adjust the switching frequency in order to obtain the suitable current and voltage gains.

523

B. Closed-Loop Control Scheme of the Proposed Converter

525

Once the resonant tanks are designed suitably according to the design procedure presented in the Section IV, the CC and CV mode charge can be simply implemented with a fixed switching frequency and guaranteed the full soft-switching condition. However, it is only possible under the assumption that the input voltage is always constant and there is no tolerance of resonant components. Hence, the operation of the proposed converter deviates from the perfect resonant condition slightly because the input voltage varies in a range of ±5%. A closed-loop control is necessary to implement the CC and CV mode charge operation properly. Therefore, a simple PI control scheme is adopted as shown in Fig. 14. The controller is composed of a PI controller, a mode transfer switch, and a limiter. An anti-wind-up scheme is adopted to reduce the overshoot and transient time caused by the limiter operation.

522

524

526 527 528 529 530 531 532 533 534 535 536 537 538 539 540

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TABLE IV DESIGN PARAMETERS OF THE PROPOSED CONVERTER Parameter

Value

EP

ro

Fig. 15. Bode plots of the proposed converter in CC mode operation under varying load conditions.

50 kHz IPW65R041CFD 1:0.55 13.2 μH 345μH 730 nF PQ50/50 1:0.55 128 μH 272 μH 25 nF 54 nF PQ50/50 HFA50PA60C

of

Switching frequency (fs ) Primary switches (S 1 –S 4 ) Turns ratio of TR 1 (1 : n 1 ) Leakage inductance of TR 1 (L lk 1 ) Magnetizing inductance of TR 1 (L m 1 ) Resonant capacitor of FBLLC1 (C r 1 ) Core size of TR 1 Turn ratio of TR 2 (1 : n 2 ) Leakage inductance of TR 2 (L lk 2 ) Magnetizing inductance of TR 2 (L m 2 ) Resonant capacitor of FBLLC2 (C r 2 ) Resonant capacitor of FBLLC2 (C r 3 ) Core size of TR 2 Rectifier diodes (D 1 –D 8 )

Fig. 17.

Waveforms of the switch S 1 .

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CV mode operation is 0.04 and 12, respectively.

Fig. 16. Bode plots of the proposed converter in CV mode operation under varying load conditions.

541 542 543 544 545 546 547 548 549 550 551 552 553 554 555

When the battery voltage is lower than the maximum charge voltage, 420 V in this research, the current controller is activated to charge the battery with a constant current. Once the voltage of the battery reaches its maximum charge voltage (420 V), the controller automatically switches its mode to the CV mode charge by turning ON the switch S M o de and by activating the voltage controller. In order to design the PI controllers, open-loop transfer functions for CC mode and CV mode are obtained by using the ac sweep function in PSIM software as shown in Figs. 15 and 16, respectively. Based on the obtained bode plots, PI controllers for the CC mode and the CV mode have been designed as shown in (40) and (41), respectively. The Kp value and Ki value of the current controller for CC mode operation is 0.02 and 2, respectively. The Kp value and Ki value of the voltage controller for

2 s 12 Gcv (s) = 0.04 + . s Gci (s) = 0.02 +

556

(40) (41) 557

VI. EXPERIMENTAL RESULTS

558

A 3.3-kW prototype battery charger has been implemented to validate the performance of the proposed dual FBLLC resonant converter. The design parameters are shown in Table IV.

559

561

A. CC Mode Operation

562

Figs. 17–21 shows the key waveforms of the proposed converter at each part during the CC mode operation. (VDC = 400 V, Vo = 250 V, Po = 1.95 kW, Io = 7.8 A, and fs = 50 kHz). Fig. 17 shows the voltage and current waveforms of the primary switch S1 . They clearly show that the ZVS turn-on and

560

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VU AND CHOI: NOVEL DUAL FB LLC RESONANT CONVERTER FOR CC AND CV CHARGES OF BATTERIES FOR ELECTRIC VEHICLES (JULY 2017)

Primary side waveforms of TR 1 and TR 2 .

of

Fig. 18.

11

Secondary side waveforms of the transformers TR 1 and TR 2 .

EP

Fig. 19.

ro

Fig. 22. Efficiency plots in CC mode operation of the proposed converter and conventional FBLLC resonant converter.

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Fig. 23. Waveforms of the switch S 1 at light load condition (V D C = 400 V, V o = 420 V, P o = 660 W, and fs = 50 kHz).

Fig. 20.

Fig. 21.

Waveforms of the diodes D 1 and D 5 .

Output current and output voltage in CC mode operation.

nearly ZCS turn-off of the primary switches are achieved in the CC charge operation. Primary and secondary side waveforms of the transformer TR1 and TR2 are shown in Figs. 18 and 19, respectively. The rms value of the primary current ipri2 RM S is 5 A as shown in Fig. 18, and the leakage inductance of the transformer TR2 (Llk2 ) is 128 μH as shown in Table IV. It can be seen that the rms value of the primary current ipri2 RM S is successfully limited by the design in step 2 in Section IV. Fig. 20 shows that the rectifier diodes D1 and D5 achieve ZCS turn-off and no reverse recovery. Fig. 21 shows the output current and voltage waveforms of the proposed converter. An efficiency plot of the proposed converter in the CC charge operation is shown in Fig. 22. This plot shows that a maximum efficiency of 98% was achieved. It also shows that the proposed converter has a higher efficiency when compared to the conventional FBLLC converter due its superior soft-switching performance.

569 570 571 572 573 574 575 576 577 578 579 580 581 582 583 584 585 586

B. CV Mode Operation

587

Figs. 23 and 24 show the waveforms of the primary switch S1 in the CV charge operation under light and heavy load

589

588

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ro

Fig. 24. Waveforms of the switch S 1 at heavy load condition (V D C = 400 V, V o = 420 V, P o = 3.3 kW, and fs = 50 kHz).

Output current and output voltage in CV mode operation.

of

Fig. 27.

Primary side waveforms of the transformers TR 1 and TR 2 .

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Fig. 25.

EP

Fig. 28. Efficiency plots in CV charge operation of the proposed converter and conventional FBLLC resonant converter.

Fig. 26.

590 591 592 593 594 595 596 597 598

Waveforms of the diodes D 1 and D 5 .

conditions, respectively. Under both conditions, the switch S1 achieves ZVS turn-on, but the turn-off current is relatively large. Fig. 25 shows waveforms of the primary sides of the transformers in both the FBLLC1 and FBLLC2 converters of the proposed converter during the CV charge operation. It is shown that both the resonant converters operate under perfect resonant conditions. The waveforms of the rectifier diodes D1 and D5 are shown in Fig. 26. Both the rectifier diodes achieve ZCS turn-off. As a result, there is no reverse recovery loss. Fig. 27 shows the

output current and output voltage waveforms of the proposed converter. It should be noted that the loss at the switch S M o de is negligible since there is only a small conduction loss (0.8 W in this case) and no switching loss during the CV mode operation. An efficiency plot of the proposed converter in the CV charge operation is shown in Fig. 28. It shows that a maximum of 97% efficiency was achieved. It also shows that the proposed converter exhibits a slightly higher efficiency when compared to the conventional FBLLC converter. C. Comparison of the Proposed Converter With Other Topologies In order to prove the superiority of the proposed converter, a comparison has been made between the proposed converter and the other conventional LLC resonant converters. Table V provides a comprehensive comparison of the proposed converter with other topologies in relation to the component counts, the complexity of control, and the soft-switching characteristics. A loss comparison between the proposed converter and the conventional FBLLC resonant converter designed with same specification is given in Fig. 29. The losses of the conventional FBLLC converter and the proposed converter are calculated at the end of CC mode charge (VDC = 400 V, Vo = 420 V, Po = 3.3 kW and Io = 7.8 A). As shown in Table V and Fig. 29, the proposed converter has many advantages in terms of losses though the component count of the proposed converter is larger than those of the other topologies.

599 600 601 602 603 604 605 606 607

608 609 610 611 612 613 614 615 616 617 618 619 620 621 622 623 624

VU AND CHOI: NOVEL DUAL FB LLC RESONANT CONVERTER FOR CC AND CV CHARGES OF BATTERIES FOR ELECTRIC VEHICLES (JULY 2017)

TABLE V COMPARISON OF THE PROPOSED CONVERTER WITH OTHER TOPOLOGIES Conventional LLC [1], [2]

References [20], [21] Fixed

Fixed

Wide range

Number of additional switches

1

0

2

Switching loss of additional switches Control scheme of additional switches Number of the resonant tanks

None

None

High

Simple

None

Complex

2

1

1

Number of the transformers Switching-on of the primary switches in CC mode Switching-off of the primary switches in CC mode Circulating current in CC mode

2

1

1

ZVS

ZVS

ZVS

Nearly ZCS High turn-off High turn-off current current Small Large Medium

[1] R. Beiranvand, B. Rashidian, M. R. Zolghadri, and S. M. H. Alavi, “Using LLC resonant converter for designing wide-range voltage source,” IEEE Trans. Ind. Electron., vol. 58, no. 5, pp. 1746–1756, May 2011, doi: 10.1109/TIE.2010.2052537. [2] F. Musavi, M. Craciun, D. S. Gautam, W. Eberle, and W. G. Dunford, “An LLC resonant DC-DC converter for wide output voltage range battery charging applications,” IEEE Trans. Power Electron., vol. 28, no. 12, pp. 5437–5445, Dec. 2013, doi: 10.1109/TPEL.2013.2241792. [3] R. Beiranvand, B. Rashidian, M. R. Zolghadri, and S. M. H. Alavi, “A design procedure for optimizing the LLC resonant converter as a wide output range voltage source,” IEEE Trans. Power Electron., vol. 27, no. 8, pp. 3749–3763, Aug. 2012, doi: 10.1109/TPEL.2012.2187801. [4] J. Deng, S. Li, C. C. Mi, and R. Ma, “Design methodology of LLC resonant converter for electric vehicle battery chargers,” IEEE Trans. Veh. Technol., vol. 63, no. 4, pp. 1581–1592, May 2014, doi: 10.1109/TVT.2013.2287379. [5] H. P. Park and J. H. Jung, “Modeling and feedback control of LLC resonant converter at high switching frequency,” J. Power Electron., vol. 16, no. 3, pp. 849–860, 2016. [6] B. Yang, F. C. Lee, A. J. Zhang, and G. Huang, “LLC resonant converter for front end DC/DC conversion,” in Proc. IEEE 17th Annu. IEEE Appl. Power Electron. Conf. Expo., 2002, vol. 2, pp. 1108–1112, doi: 10.1109/APEC.2002.989382. [7] C. W. Tsang, M. P. Foster, D. A. Stone, and D. T. Gladwin, “Analysis and design of LLC resonant converters with capacitor-diode clamp current limiting,” IEEE Trans. Power Electron., vol. 30, no. 3, pp. 1345–1355, Mar. 2015, doi: 10.1109/TPEL.2014.2317653. [8] K. H. Yi, and G. W. Moon, “Novel two-phase interleaved LLC seriesresonant converter using a phase of the resonant capacitor,” IEEE Trans. Ind. Electron., vol. 56, no. 5, pp. 1815–1819, May 2009, doi: 10.1109/TIE.2008.2011310. [9] Z. Hu, Y. Qiu, Y. F. Liu, and P. C. Sen, “A control strategy and design method for interleaved LLC converters operating at variable switching frequency,” IEEE Trans. Power Electron., vol. 29, no. 8, pp. 4426–4437, Aug. 2014, doi: 10.1109/TPEL.2014.2300165. [10] H. Wang, S. Dusmez, and A. Khaligh, “Maximum efficiency point tracking technique for LLC-based PEV chargers through variable DC link control,” IEEE Trans. Ind. Electron., vol. 61, no. 11, pp. 6041–6049, Nov. 2014, doi: 10.1109/TIE.2014.2311399. [11] W. Feng, F. C. Lee, and P. Mattavelli, “Simplified optimal trajectory control (SOTC) for LLC resonant converters,” IEEE Trans. Power Electron., vol. 28, no. 5, pp. 2415–2426, May 2013, doi: 10.1109/ TPEL.2012.2212213. [12] R. Beiranvand, B. Rashidian, M. R. Zolghadri, and S. M. H. Alavi, “Optimizing the normalized dead-time and maximum switching frequency of a wide-adjustable-range LLC resonant converter,” IEEE Trans. Power Electron., vol. 26, no. 2, pp. 462–472, Feb. 2011, doi: 10.1109/TPEL.2010.2068563. [13] Z. Fang, T. Cai, S. Duan, and C. Chen, “Optimal design methodology for resonant converter in battery charging applications based on time-weighted average efficiency,” IEEE Trans. Power Electron., vol. 30, no. 10, pp. 5469–5483, Oct. 2015, doi: 10.1109/TPEL.2014.2379278. [14] Z. G. Liang, R. Guo, G. Y. Wang, and A. Huang, “A new wide input range high efficiency photovoltaic inverter,” in Proc. IEEE Energy Convers. Congr. Expo., 2010, pp. 2937–2943, doi: 10.1109/ECCE.2010.5618217. [15] J. H. Kim, C. E. Kim, J. K. Kim, J. B. Lee, and G. W. Moon, “Analysis on load-adaptive phase-shift control for high efficiency full-bridge resonant converter under light-load conditions,” IEEE Trans. Power Electron., vol. 31, no. 7, pp. 4942–4955, Jul. 2016, doi: 10.1109/TPEL.2015. 2462077. [16] B. McDonald and F. Wang, “LLC performance enhancements with frequency and phase shift modulation control,” in Proc. IEEE Appl. Power Electron. Conf. Expo., Mar. 2014, pp. 2036–2040., doi: 10.1109/ APEC.2014.6803586. [17] Y. K. Lo, C. Y. Lin, M. T. Hsieh, and C. Y. Lin, “Phase-shifted full-bridge series-resonant DC-DC converters for wide load variations,” EEE Trans. Ind. Electron., vol. 58, no. 6, pp. 2572–2575, Jun. 2011, doi: 10.1109/TIE.2010.2058076. [18] H. Wu, T. Mu, X. Gao, and Y. Xing, “A secondary-side phaseshift-controlled resonant converter with reduced conduction loss at normal operation for hold-up time compensation application,” IEEE Trans. Power Electron., vol. 30, no. 10, pp. 5352–5357, Oct. 2015, doi: 10.1109/TPEL.2015.2418786.

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Reverse recovery loss of rectifier None Exist None diodes in CC mode Switching-on of the primary ZVS ZVS ZVS switches in CV mode Switching-off of the primary High turn-off High turn-off High turn-off switches in CV mode current current current Circulating current in CV mode Large Large Medium

REFERENCES

of

Switching frequency

Proposed

625 626 627 628 629 630 631 632 633 634 635 636 637 638 639 640

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Fig. 29. Loss comparison between the proposed converter and conventional FBLLC converter at full load condition.

VII. CONCLUSION

This paper introduced a novel dual FBLLC resonant converter suitable for implementing the CC and CV charge of the batteries. By designing the two resonant converters to resonate at lower and higher resonant frequencies, respectively, it is possible to obtain the advantages of each operation, which results in ZVS and nearly ZCS of the primary switches in the CC charge operation. The CV charge is implemented by operating both the converters at the higher resonant frequency by transforming the resonant tank of the FBLLC2 resonant converter. The converter operates with no switching losses at the primary switches and no switching frequency variation over the entire charge process. Its high efficiency over a wide range of output power makes it suitable for battery charge applications, such as electric vehicles and energy storage systems.

13

641 642 643 644 645 646 647 648 649 650 651 652 653 654 655 656 657 658 659 660 661 662 663 664 665 666 667 668 669 670 671 672 673 674 675 676 677 678 679 680 681 682 683 684 685 686 687 688 689 690 691 692 693 694 695 696 697 698 699 700 701 702 703 704 705 706 707 708 709 710 711 712 713 714

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Hai-Nam Vu received the B.S. degree from the Hanoi University of Science and Technology, Hanoi, Vietnam, in 2014, and the M.S. degree from Soongsil University, Seoul, South Korea, in 2017, both in electrical engineering. From 2014 to 2015, he was an Assistant Researcher in the Centre for Technology Innovation, Hanoi University of Science and Technology. His research interests include dc–dc converters dealing with the renewable sources and batteries for energy storage system or hybrid electric vehicles, soft-switching techniques for the pulse width modulation/pulse-frequency modulation converters, and resonant converters.

Woojin Choi (S’00–M’05) was born in Seoul, South Korea, in 1967. He received the B.S. and M.S. degrees from Soongsil University, Seoul, South Korea, in 1990 and 1995, respectively, and the Ph.D. degree from Texas A&M University, College Station, TX, USA, in 2004, all in electrical engineering. From 1995 to 1998, he was a Research Engineer in Daewoo Heavy Industries, Seoul, South Korea. In 2005, he joined the School of Electrical Engineering, Soongsil University, Seoul, South Korea. His research interests include modeling and control of electrochemical energy sources, such as fuel cells, batteries, and supercapacitors; power conditioning technologies in renewable energy systems; and dc–dc converters for electric vehicles and fuel cells. Dr. Choi is an Associate Editor of the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS and a Publication Editor of the Journal of Power Electronics of the Korean Institute of Power Electronics.

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[19] K. Zheng, D. Zhou, J. Li, L. Li, and Y. Zhao, “A digital self-sustained phase shift modulation control strategy for full-bridge LLC resonant converters,” J. Power Electron., vol. 16, no. 3, pp. 915–924, 2016. [20] X. Sun, X. Li, Y. Shen, B. Wang, and X. Guo, “A modified high-efficiency LLC converter with two transformers for wide input-voltage range application,” IEEE Trans. Power Electron., vol. 28, no. 4, pp. 1946–1960, 2016, doi: 10.1109/TPEL.2012.2201959. [21] H. Huau, X. Fang, F. Chen, Z. J. Shen, and I. Batarseh, “A dual-bridge LLC resonant converter with fixed-frequency PWM control for wide input applications,” IEEE Trans. Power Electron., vol. 28, no. 1, pp. 1946–1960, Apr. 2013, doi: 10.1109/TPEL.2016.2530748. [22] J. Dodge, “Power MOSFET tutorial,” Adv. Power Technol., Beach, FL, USA, Appl. Note APT0103.

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743 744 745 746 747 748 749 750 751 752 753 754 755 756 757 758 759 760 761

Queries

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Q1. Author: Please confirm if “(July 2017)” should be present in the article title. Q2. Author: Please provide the expanded form of “CC,” “CV,” “ZCS,” and “ZVS” at their first occurrence in the text. Q3. Author: Please provide the pincode of “Hanoi University of Science and Technology” in the affiliation of H.-N. Vu. Also, Please check if the affiliations are okay as set for correctness. Q4. Author: Please check if (19) should be replaced with (20) in the sentence starting rom “Since the FBLLC2 resonant converter operates . . . .” Q5. Author: The captions of Figs. 20 and 26 are similar. Please check. Q6. Author: Please provide year in Ref. [22].

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A Novel Dual Full-Bridge LLC Resonant Converter for CC and CV Charges of Batteries for Electric Vehicles (July 2017) Hai-Nam Vu and Woojin Choi, Member, IEEE

4

24 25 26

Index Terms—CC/CV charge, dual full-bridge LLC (FBLLC) resonant converter, high efficiency, variable resonant network.

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I. INTRODUCTION

28

LECTRIC vehicles (EVs) and Plug-in hybrid electric vehicles (PHEVs) are becoming more popular in an effort to guarantee energy security and to lower fuel consumption and emissions. On-board chargers (OBCs) with high power density and high power conversion efficiency are crucial for the proliferation of EVs and PHEVs. An OBC is normally composed of an ac–dc converter followed by a dc–dc converter to charge batteries from a dc link. One of the most popular dc–dc converters for battery chargers is the full-bridge LLC (FBLLC) resonant converter. However, the FBLLC resonant converter needs to utilize one of the three control methods, such as frequency

29 30 31 32 33 34 35 36 37 38

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modulation, phase-shift modulation, or a modified structure in order to implement CC and CV charges for batteries. In [1]–[13], frequency modulation methods are used to control the output voltage or output current of an FBLLC resonant converter. The LLC resonant converter with frequency modulation has been researched in many aspects. The research works in [1]–[5] focus on the design of the LLC resonant converter for a wide output voltage range such as the battery charge applications. In [6], it is suggested to use the LLC resonant converter for the distributed power system operating at a high switching frequency due to its ZVS turn-on characteristics. In [7], a design methodology for the LLC resonant converters by using a capacitor-diode clamp is proposed to limit the current in the overload conditions. For the high-power applications, an interleaved LLC resonant converter is suggested to reduce the output voltage ripple [8] and to limit the effects of the component tolerances [9]. Some methods to track the maximum efficiency point of the LLC resonant converter based on the state-of-charge of a battery are introduced in [10] and [11]. In [12] and [13], optimal design of LLC resonant converter is discussed in relation to the dead time, the maximum switching frequency, and the time-weighted average efficiency. However, all of the methods mentioned in [1]–[13] are not able to implement the CC/CV charge of the battery without the wide range of the switching frequency variation. Once the resonant converter operates out of the resonant frequency, it loses its main advantages such as a low switching loss and a low circulating current. As a result, it becomes less efficient. When the switching frequency is higher than the resonant frequency, the primary switches of the FBLLC resonant converter suffer from a high turn-off current. When the switching frequency is lower than the resonant frequency, the circulating current occurs in the primary side. In addition, the wide range of the switching frequency variation causes difficulty in optimizing the magnetic components [14]. Besides, higher conduction losses and a lower power density of the converter are unavoidable when it is not optimized. In [15]–[19], a phase-shift modulation method is used to increase the efficiency of an FBLLC resonant converter under a light load condition. Compared to the frequency modulation method, phase-shift modulation can reduce the peak-to-peak flux density of the transformer. As a result, the core loss of the transformer can be significantly reduced and a high conversion efficiency can be achieved at a light load. However, the

ro

7 8 9 10 11 12 13 14 15 16 17 18 19 20 21 22 23

Abstract—This paper introduces a novel concept for implementing CC and CV charges of batteries for electric vehicles using a dual full-bridge LLC (FBLLC) resonant converter, which shares its primary switches. In the proposed concept, the CC and CV charges are implemented using the inherent characteristics of an LLC converter as a current source and a voltage source. One FBLLC resonant converter operates with a fixed-resonant network whereas the other operates with a variable resonant network for the CC and CV charge operations. The proposed converter can achieve ZVS and nearly ZCS for all of the primary switches in the CC charge operation, and ZVS for all of the primary switches in the CV charge operation, which lead to a high efficiency. In addition, fixed switching frequency operation is possible in both the CC and CV charges thanks to the variable resonant network. A 3.3-kW prototype converter is implemented to verify the feasibility and validity of the proposed converter and a maximum efficiency of 98% was achieved.

5

Q1 6

of

1

1

Manuscript received February 15, 2017; revised May 1, 2017 and June 26, 2017; accepted July 12, 2017. (Corresponding author: Woojin Choi.) H.-N. Vu was with the Department of Electrical Engineering, Soongsil University, Seoul 156-743, South Korea. He is now with the Centre for Technology Innovation, Hanoi University of Science and Technology, Hanoi, Vietnam (e-mail: [email protected]). W. Choi is with the Department of Electrical Engineering, Soongsil University, Seoul 156-743, South Korea (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIE.2017.2739705

0278-0046 © 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications standards/publications/rights/index.html for more information.

39 40 41 42 43 44 45 46 47 48 49 50 51 52 53 54 55 56 57 58 59 60 61 62 63 64 65 66 67 68 69 70 71 72 73 74 75 76 77 78 79 80 81

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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS

Fig. 2.

86 87 88 89 90 91 92 93 94 95 96 97 98 99 100 101 102 103 104 105 106 107 108 109 110 111

112 113 114 115 116 117 118 119 120 121 122 123 124 125 126

Vo (s) = n

1+

Vac

L r s+ C 1r s Lm s

 + Rac Lr s +

1 Cr s



fr 2 = Io (s) =







n Lr s +

1 Cr s

 1 + Rac

132 133 134 135 136 137 138 139

140 141 142 143 144 145 146 147 148 149

(3)

(Lm + Lr )Cr .

130 131

(2)

1 Vin

129

.

The lower resonant frequency fr 2 is determined by the resonant components Lr and Cr plus the magnetizing inductance Lm , as shown in (3). When the FBLLC resonant converter operates at fr 2 , its output current is independent of the load variation since the term associated with the load resistance Rac becomes zero in the denominator as shown in (4). The FBLLC resonant converter works as a constant current source at fr 2 . However, the lower resonant frequency is rarely used since the converter can achieve ZCS turn-off but loses ZVS turn-on, which is not preferred for power MOSFETs [22].

II. PROPOSED DUAL FBLLC RESONANT CONVERTER

It is well known that the LLC resonant converter can operate as a voltage source or as a current source at certain resonant frequencies. The proposed dual FBLLC resonant converter (see Fig. 1) utilizes these inherent characteristics to implement the CC and CV charges. The CC charge can be implemented by operating the upper and lower FBLLC converters as a voltage and current source, respectively. The CV charge can be implemented by operating both of the converters as voltage sources. The basic characteristics of the FBLLC converter are summarized in the Section II-A. The structure and operating principle of the proposed converter are described in Section II-B. Sections II-C and II-D show the methods for implementing the CC and CV charge operations and designing the resonant tanks of the proposed converter, respectively.

127 128

Fig. 2 shows an equivalent circuit of a single FBLLC resonant converter that was derived by first harmonic approximation, and it has two resonant frequencies [2], [3]. The higher resonant frequency fr 1 is determined by the resonant components Lr and Cr , as in (1). When the FBLLC resonant converter operates at fr 1 , its voltage gain is independent of the load variation since the term associated with the load resistance Rac becomes zero in the denominator as shown in (2) [2]. Hence, the FBLLC resonant converter works as a constant voltage source at fr 1 , and the primary switches can achieve the ZVS turn-on 1 √ fr 1 = (1) 2π Lr Cr

ro

85

A. Two Resonant Frequencies of a Single FBLLC Resonant Converter

EP

83 84

phase-shift modulation method is not preferred for heavy loads due to its high turn-off currents for the primary switches when compared with the frequency modulation method. In order to avoid the above-mentioned problems, modified structures of an FBLLC resonant converter are presented in [20]–[21]. These FBLLC resonant converters can transform their structures from full-bridge to half-bridge and vice versa by using additional switches. Hence, it is possible to regulate the output voltage with a minimal operating frequency variation and to optimize the design of the magnetic components. However, these kinds of converters require two additional active switches, which operate with hard switching, to transform their structure and all of the active switches suffer from high turn-off currents leading to a lower efficiency. In this paper, a novel dual FBLLC resonant converter is proposed to cope with all the aforementioned problems as shown in Fig. 1. The advantages of the proposed converter can be summarized as follows: 1) ZVS and nearly ZCS for all primary switches in CC charge, and ZVS for all primary switches in CV charge; 2) Only one additional switch to transform the resonant tank and no switching loss associated with it; 3) A fixed-frequency resonant operation in both CC and CV charge and hence lower circulating energy; 4) Perfect soft switching of all the rectifier diodes due to the resonant operation over the entire CC and CV mode charge, hence no reverse recovery problems; and 5) High voltage gain and lower voltage rating of the rectifier diodes due to the series connection of two transformers and two rectifier bridges.

IEE

82

Proposed dual full-bridge LLC resonant converter.

of

Fig. 1.

AC equivalent circuit of a single FBLLC converter.

1 Lm s L r s+ C 1r s

 +1 .

(4)

B. Structure and Operating Principle of the Proposed Converter The above-mentioned characteristics of the FBLLC converter can be utilized to implement the CC and CV charges of the OBC of EVs. The proposed converter is composed of two FBLLC converters, which share primary switches as shown in Fig. 1. In the proposed converter, the primary switches always operate with a fixed frequency and a fixed duty cyle of 50%. Whereas the FBLLC1 converter is designed to always work as a voltage source at the higher resonant frequency, and the FBLLC2 converter has a variable structure with an additional switch to

150 151 152 153 154 155 156 157 158 159 160

VU AND CHOI: NOVEL DUAL FB LLC RESONANT CONVERTER FOR CC AND CV CHARGES OF BATTERIES FOR ELECTRIC VEHICLES (JULY 2017)

3

TABLE I RESONANT FREQUENCIES USED TO IMPLEMENT THE CC AND CV CHARGE OF THE PROPOSED CONVERTER CC mode charge

CV mode charge

Used

Used

Not Used

Not Used

Not Used

Not Used

fr 2 L L C 2 = 1 √

Used

Not Used

L lk 2 , L m 2 , fr 1 L L C 3 = 1 √ (C r 2 + C r 3 )

Not Used

Used

Not Used

Not Used

L lk 1 , L m 1 , fr 1 Cr 1

LLC1

=





1 L lk 1 C r 1

fr 2 L L C 1 = 1 √ 2π

FBLLC2 converter

L lk 2 , L m 2 , fr 1 Cr 2





(L m 1 + L l k 1 )C r 1

LLC2

=

L l k 2 (C r 2 + C r 3 )

164 165 166 167 168 169 170 171 172 173 174 175 176 177 178 179 180 181 182 183 184

185 186 187 188 189 190 191 192 193

1

(L m 2 + L l k 2 )(C r 2 + C r 3 )

Fig. 3. AC equivalent circuit of the proposed dual FBLLC resonant converter in the CC mode operation.

transform its resonant tank, and hence the resonant frequency, to make it work as a current source for CC charge and as a voltage source for CV charge. Table I shows three different resonant tanks formed in the proposed dual FBLLC resonant converter, their resonant frequencies, and their usages in implementing the CC and CV charge. The upper converter has a resonant tank composed of Llk1 , Lm 1 , and Cr 1 . Only the resonant operation at the higher resonant frequency fr 1 LLC1 is used for both the CC and CV charges. The lower converter has two resonant tanks, one composed of Llk2 , Lm 2 , and Cr 2 , and the other composed of Llk2 , Lm 2 , and (Cr 2 + Cr 3 ). In the CC charge, the lower converter operates at the lower resonant frequency fr 2 LLC2 so that it can work as a current source. In the CV charge, it operates at the higher resonant frequency fr 1 LLC3 so that it can work as a voltage source by turning ON an additional switch S M o de . Here, the resonant network of the FBLLC2 converter is designed to have a lower resonant frequency fr 2 LLC2 , which is the same as the higher resonant frequency fr 1 LLC1 of the upper resonant converter. In the CV charge, both the upper and lower converters work as a voltage source by the resonant operation at fr 1 LLC1 and fr 1 LLC3 , respectively. In this mode, an additional switch is turned ON to transform the resonant tank of the FBLLC2 converter to resonate at fr 1 LLC3 .

EP

163

L lk 2 C r 2

IEE

161 162



(L m 2 + L l k 2 )C r 2

fr 2 L L C 3 = √ 2π



1

of

FB LLC1 converter

Higher and lower resonant frequencies of each resonant tank

ro

Resonant tank

C. Design Considerations for the CC Mode Operation of the Proposed Converter As previously mentioned, the CC charge can be implemented by operating the FBLLC1 resonant converter as a voltage source and by operating the FBLLC2 resonant converter as a current source. Hence, the FBLLC1 resonant converter and the FBLLC2 resonant converter need to resonate at fr 1 LLC1 and fr 2 LLC2 , respectively. It should be noted that the FBLLC1 converter must be designed to have a lower output voltage than that of the

Fig. 4. Voltage gain of FBLLC1 and output current of FBLLC2 curves w.r.t. the switching frequency variation under different load conditions.

minimum battery voltage (250 V in this research) in order to operate the entire converter as a current source. The switch S M o de is turned OFF during the CC charge operation. In this case, the resonant tank of the FBLLC2 resonant converter is composed of Llk2 , Lm 2 , and Cr 2 . The equivalent circuit of the proposed converter in the CC charge can be drawn as shown in Fig. 3. The higher resonant frequency of the FBLLC1 resonant converter fr 1 LLC1 and the lower resonant frequency of the FBLLC1 resonant converter fr 2 LLC2 can be calculated, respectively, by the following equations: fr 1

LLC1

=

fr 2

LLC2

=

1 2π Llk1 Cr 1 2π



(5)

1  . (Lm 2 + Llk2 )Cr 2

(6)

Since the two converters share the primary switches, the resonant tank of two converters need to be designed to have the same resonant frequency, as shown in the following equation: fr 1

LLC1

= fr 2

LLC2 .

194 195 196 197 198 199 200 201 202 203 204

205 206 207

(7)

Fig. 4 shows the voltage gain curves for the FBLLC1 resonant converter and the output current curves for the FBLLC2 resonant

208 209

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IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS

212 213 214 215 216 217

converter under different load conditions. As shown in Fig. 4, the FBLLC1 converter and the FBLLC2 converter both resonate at a switching frequency fs , thereby operating the FBLLC1 resonant converter as a constant voltage source and the FBLLC2 resonant converter as a constant current source, respectively. As shown in Fig. 3, the output of the FBLLC1 resonant converter is represented by a constant voltage source Vo1 , which can be calculated by

ro

210 211

Vo1 = n1 Vin 219 220 221

where n1 is the turns ratio of the transformer TR1 of the FBLLC1 resonant converter. The output of the FBLLC2 resonant converter can be represented by a constant current source Io2 , which is calculated by Io2 =

223 224 225 226

Vin

  n2 ωs Llk2 −

1 ωs Cr 2

  .

Io = Io2 .

227 228

230 231 232 233 234 235 236 237 238 239 240

(10)

Therefore, the output power of the FBLLC1 resonant converter and FBLLC2 resonant converter can be represented by PFBLLC1 = Vo1 Io , PFBLLC2 = (Vo − Vo1 )Io .

229

(9)

In the CC charge, the output of the proposed converter is composed of a voltage source and a current source connected in series. The output current of the FBLLC2 converter is the same as the charge current of the battery in the CC charge as shown in the following equation:

IEE

222

(8)

Fig. 6. Voltage gain curves of the FBLLC1 and the FBLLC2 converters w.r.t. the switching frequency variation under different load conditions.

FBLLC2 resonant converter can be expressed by fr 1

EP

218

of

Fig. 5. AC equivalent circuit of proposed converter in CV charge operation.

(11)

D. Design Considerations for the CV Mode Operation of the Proposed Converter In order to implement the CV charge of the proposed converter, both converters work as voltage sources. Since the resonant tank of the FBLLC2 used for the CC charge is designed to satisfy (7), the higher resonant frequency of it fr 1 LLC2 = √ 1 cannot be the same as the higher resonant frequency 2π L l k 2 C r 2 of the FBLLC1 resonant converter fr 1 LLC1 . Hence, the switch S M o de is turned ON to transform the resonant tank in this mode. Since the resonant capacitor Cr 2 and Cr 3 are connected in parallel, the total capacitance increases to (Cr 2 + Cr 3 ) as shown in Fig. 5. Hence, in this mode, the resonant frequency of the

LLC3

=

1  . 2π Llk2 (Cr 2 + Cr 3 )

241

(12) 242

Since both the converters have to resonate at the same switching frequency in the CV charge operation, the higher resonant frequency of the FBLLC2 resonant converter fr 1 LLC3 needs to be designed to have the same resonant frequency as the higher resonant frequency of the FBLLC1 resonant converter fr 1 LLC1 as shown in the following equation: fr 1

LLC3

= fr 1

LLC1 .

246 247 248

249 250 251 252 253 254 255

(14)

where n2 is the turns ratio of the transformer TR2 of the FBLLC2 resonant converter. In this CV charge, the output of the proposed converter is composed of two voltage sources connected in series and the output voltage can be represented by Vo = Vo1 + Vo2 = Vin (n1 + n2 ) .

244 245

(13)

Fig. 6 shows the voltage gain curves for the FBLLC1 resonant converter and the FBLLC2 resonant converter under different load conditions. As shown in Fig. 6, the FBLLC1 converter and the FBLLC2 converter both resonate at a switching frequency fs , thereby operating both of them as constant voltage sources. The output of the FBLLC2 resonant converter can be represented by a constant voltage source Vo2 which is calculated by Vo2 = n2 Vin .

243

256 257 258 259

(15)

E. Charge Sequence of the Proposed Converter

260

The specification of the proposed converter is shown in Table II, and the CC/CV charge profile of the EV battery and its equivalent impedance during the charge are shown in Fig. 7. The battery pack is composed of 100 Li-ion battery cells and

261 262 263 264

VU AND CHOI: NOVEL DUAL FB LLC RESONANT CONVERTER FOR CC AND CV CHARGES OF BATTERIES FOR ELECTRIC VEHICLES (JULY 2017)

5

TABLE II RESONANT SPECIFICATION OF THE PROPOSED CONVERTER Parameter Input voltage Output voltage (battery voltage) Power rating CC charge current CV charge voltage Resonant frequency

Designator

Value

V in Vo Po Io Vo fr

380–420 V 250–420 V 3.3 kW 7.8 A 420 V 50 kHz

of

Fig. 8. Power distribution FBLLC1 and FBLLC2 resonant converter in CC and CV mode operation.

268 269 270 271 272 273 274 275 276 277 278 279 280 281

282 283 284 285 286 287 288 289 290 291 292 293 294

III. MODES OF OPERATION OF THE PROPOSED DUAL FBLLC RESONANT CONVERTER

300

In this section, the modes of operation of the proposed converter in the CC and CV charges are detailed. Some of the circuit parameters are predefined as follows. The angular switching frequency and the switching period are expressed by (16) and (17), respectively

EP

267

the voltage of it varies from 250 to 420 V. The charge starts with CC mode charge with a constant current of 7.8 A. The CC mode charge is finished when the voltage of the battery pack reaches its maximum voltage (420 V). After then, the proposed converter switches its charge mode to CV mode charge by turning ON the switch S M o de . The CV mode is ended when the charge current is decreased to 0.78 A (0.1 C). Here, the output voltage of the FBLLC1 converter is designed to take a half of the maximum battery voltage (210 V), which is less than the minimum battery voltage (250 V). Therefore, in the CC mode the charge current depends solely on the output current of the FBLLC2 resonant converter. The FBLLC1 converter must be designed to have a lower output voltage than that of the minimum battery voltage (250 V) in order to operate the entire converter as a current source. Otherwise constant current mode cannot be implemented since the FBLLC1 becomes active due to its higher voltage than that of the battery.

IEE

266

298

ro

Fig. 7. CC/CV charge profile of the EV battery pack and its equivalent impedance during the charge.

265

voltages have the same values. Since the maximum output powers of both resonant converters are the same, the components with same ratings can be used for each converter, which is advantageous.

F. Power Distribution in FBLLC1 Resonant Converter and FBLLC2 Resonant Converter Since the output of two FBLLC resonant converters is connected in series, the output power of each FBLLC resonant converter is proportional to its output voltages in both CC and CV mode operations. The power distribution in two converters during CC and CV mode operations is shown in Fig. 8. During the CC mode charge operation, the output power of FBLLC1 is 1638[W] (50% of the total output power) and the output power of FBLLC2 increases gradually from 312[W] (19% of the total output power) to 1638[W] (50% of the total output power). In the CV mode operation, the output powers of FBLLC1 and FBLLC2 are equal when their output

ωs = 2πfs Ts =

1 . fs

295 296 297

299

301 302 303 304 305

(16) (17)

A. CC Charge Operation

306

The switch S M o de is always turned OFF during the CC charge operation. There are four operating modes in a half switching cycle of the proposed converter in the CC charge operation. The key waveforms for each of the operating modes are illustrated in Fig. 9. Mode 1 (t0 –t1 ): Switches S2 and S3 are turned OFF with nearly ZCS at t0 . The primary current of the FBLLC resonant converter ipri discharges the output capacitance of the switches S1 and S4 . After the voltage of the switches S1 and S4 drops to zero, the body diodes of these switches conduct and achieves zero-voltage switching conditions. The slope of the magnetizing current im 1 (t) can be expressed by

307

dim 1 (t) Vin = . dt Lm 1

4Vin sin (ωS (t − t1 ) − θpri1 ) π |Zin1 (jωS )|

309 310 311 312 313 314 315 316 317 318

(18)

Mode 2 (t1 –t2 ): The switches S1 and S4 are turned ON with ZVS at t1 . The resonant tank of the FBLLC1 resonant converter, which is composed of Llk1 and Cr 1 , starts to resonate and the primary current ipri1 (t) can be expressed by ipri1 (t) =

308

(19)

319 320 321 322

6

IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS

Therefore, the summation of the above two primary currents expressed by (19) and (20) is the same as the primary switch current ids1 as shown in Fig. 7. Mode 3 (t2 –t3 ): At t2 , the magnitude of the primary current ipri2 (t2 ) is equal to that of the magnetizing current of the transformer TR2 in the FBLLC2 resonant converter. The power transfer through it ends and the rectifier diode D5 achieves ZCS turn-off. In this mode, the primary currents ipri1 and ipri2 can be expressed by (19) and (20), respectively. The magnetizing current im 2 (t) can be expressed by Vo2 (t − t2 ) . n2 Lm 2

326 327 328 329 330 331

Q4

332

N =1,3,5...

4Vin sin (N ωS (t − t1 ) N π |Zin2 (jN ωS )|

− θpri2 N ) 333 334 335 336 337

(20)

where Zin2 (jN ωS ) is the input impedance of the FBLLC2 resonant converter at the angular switching frequency N ωS , and θpri2 N is the phase angle of the N th component of the primary current ipri2 with respect to that of the N th component of the voltage vab .

343 344 345 346 347

B. CV Charge Operation

359

The switch S M o de is turned ON during the CV charge operation. In this operation, both the FBLLC1 and FBLLC2 converters work at their higher resonant frequencies. The operating principle in CV charge operation is the same as the conventional FBLLC resonant converter and it can be found in [2]. In this mode, if the converter operates at the frequency higher than 31 kHz, the converter operates in inductive region and the ZVS condition for all of the primary switches can be guaranteed regardless of the load variation.

360

368

IV. DESIGN PROCEDURE FOR RESONANT NETWORKS

369

In this section the design procedure for the resonant tanks is detailed. In order to implement the CC and CV charges with fixed-frequency operation, the resonant tanks need to be designed carefully by the following procedure. Step 1 (Design of magnetizing inductance Lm 2 ): In CC charge operation, the FBLLC2 resonant converter operates at its lower resonant frequency fr 2 LLC2 . Therefore, as shown in Fig. 3 and (6), the impedance of the resonant tank in FBLLC2 becomes zero at the angular switching frequency ωs and (22) can be derived    1   . (22) |ωs Lm 2 | = ωs Llk2 − ωs Cr 2 

370

EP

ipri2 (t) =

∞ 

342

348

ro 325

where Zin1 (jωS ) is the input impedance of the FBLLC1 resonant converter at the angular switching frequency ωS and θpri1 is the phase angle of the primary current ipri1 with respect to that of the fundamental component of the voltage vab . At the same time, the resonant tank of the FBLLC2 resonant converter, which is composed of Lm 2 , Llk2 , and Cr 2 , starts to resonate and the primary current ipri2 can be expressed by (20). Since the FBLLC2 resonant converter operates at the lower resonant frequency fr 2 LLC2 , its primary current can be expressed by the Fourier series in (19)

IEE

323 324

341

Mode 4 (t3 –t4 ): At t3 , the switches S1 and S4 are turned with nearly ZCS, and the rectifier diode D1 achieves ZCS turn-off. As mentioned in Mode 2, since the switch current is a summation of the two primary currents of the FBLLC resonant converters, some design rules should be applied to design the transformer TR1 of the FBLLC1 resonant converter to achieve nearly ZCS turn-off of the primary switches. This is described in detail in the following section. Since the operation of the proposed converter during the second half of the switching cycle is symmetric to that during the first half of the switching cycle, it is omitted here. OFF

Fig. 9. Key waveforms of the proposed converter in CC charge operation.

339 340

(21)

of

im 2 (t) = im 2 (t2 ) −

338

By using (9), (10), (14), and (22), the magnetizing inductance Lm 2 of FBLLC2 converter can be calculated by Lm 2 =

Vin . n2 ωs Io

349 350 351 352 353 354 355 356 357 358

361 362 363 364 365 366 367

371 372 373 374 375 376 377 378 379

380 381

(23)

Step 2 (Design of leakage inductance Llk2 ): In the FBLLC2 resonant converter, the rms value of the primary current ipri2

382 383

VU AND CHOI: NOVEL DUAL FB LLC RESONANT CONVERTER FOR CC AND CV CHARGES OF BATTERIES FOR ELECTRIC VEHICLES (JULY 2017)

7

the value of Llk2 . Therefore, the magnetizing inductance Llk2 should be designed to be as large as possible in order to minimize the rms values of the primary current ipri2 RM S , which results in lower conduction losses of the primary switches. A leakage inductance value larger than 120 μH is suitable to keep the rms value of the primary current ipri2 less than 5.5 A. Step 3 (Design of resonant capacitor Cr 2 ): The resonant capacitor Cr 2 can be calculated from (6) as Cr 2 =

1 ωs2

(Lm 2 + Llk2 )

.

387 388 389 390

It can be seen from (24) that the magnitude of Ipri2 RM S depends on the values of |Zin2 (jN ωS )|. The input impedance of the FBLLC2 resonant converter at the angular switching frequency N ωS can be expressed by

of

Zin2 (jN ωs ) =

1 + jN ωs Llk2 jN ωs Cr 2

+

392 393 394

n22 π82

jN ωs Lm 2 Rac2 jN ωs Lm 2 + Rac2

(25)

o2 Ro V o 1V+V o2

where Rac2 = . Since Cr 2 can be calculated by 1/(ωs 2 (Lm 2 + Llk2 )) at the lower resonant frequency, the input impedance Zin2 (jN ωS ) can be expressed by

IEE

391

Zin2 (jN ωs ) =

1 jN ωs ω 2 (L m 21+L l k 2 ) s

+ jN ωs Llk2 +

395 396 397 398 399 400 401 402 403 404

− Cr 2 .

jN ωs Lm 2 Rac2 . (26) jN ωs Lm 2 + Rac2

Equation (26) shows that the input impedance Zin2 (jN ωs ) depends on Llk2 and Lm 2 once the load impedance is designated. Since Lm 2 is already decided in step 1, the rms value of the primary current Ipri2 RM S only depends on the value of Llk2 . Fig. 10 illustrates the relationship between the leakage inductance Llk2 and the rms current value of the primary current ipri2 of the FBLLC2 resonant converter under a certain operating condition (VDC = 400 V, Vo = 250 V, Po = 1.95 kW, Io = 7.8 A, and fs = 50 kHz). As shown in Fig. 10, the rms values of the primary current ipri2 is inversely proportional to

Cr 1 =

1 ωs2 Llk1

.

411 412

413 414

415 416 417 418 419 420 421 422 423 424 425

426 427 428 429 430 431 432 433 434 435 436 437 438 439 440

(30)

The total inductive energy EL stored in the primary side at the switching moment can be calculated by EL =

409 410

(29)

Step 7 (Design of magnetizing inductance Lm 1 ): In the CC charge operation of the proposed converter, the FBLLC1 and FBLLC2 converters operate in the ZVS and ZCS regions, respectively. In order to ensure the soft switching at both the turn-on and turn-off of the switches, the proposed converter, which is a combination of two FBLLC converters, should operate in the ZVS region over the entire load range during the CC charge. In this section, the magnetizing inductance Lm 1 is designed to ensure ZVS and nearly ZCS for all of the primary switches over the entire load of the CC charge. The ZVS for all of the primary switches can be achieved by designing the inductive energy EL stored in the primary side to have a higher value than the capacitive energy EC stored in the output capacitances of the primary switches by the following equation: EL ≥ E C .

408

(28)

ro

385 386

depends on the leakage inductance Llk2 . The rms value of the primary current ipri2 can be calculated from (20) by the following equation:    ∞  4Vin √ . (24) Ipri2 RM S = 2N π |Zin2 (jN ωS )| N =1,3,5...

1

ωs2 Llk2

Step 5 (Design of leakage inductance Llk1 ): There are no specific requirements for the design of the leakage inductance Llk1 since the FBLLC1 converter always operates at its higher resonant frequency and only the magnetizing current contributes to the discharge of the output capacitors of the primary switches. However, it is better to minimize this since the voltage stress applied to the resonant capacitor Cr 1 is increased when the value of the leakage inductance Llk1 is large. Step 6 (Design of resonant capacitor Cr 1 ): The resonant capacitor Cr 1 can be calculated using (29) after determining the leakage inductance Llk1 in the previous step

EP

384

Cr 3 =

RMS value of the primary current ip ri2 w.r.t. L lk 2 value.

406 407

(27)

Step 4 (Design of resonant capacitor Cr 3 ): The resonant capacitor Cr 3 can be calculated from (12) as Fig. 10.

405

441 442

1 1 1 Lm 1 i2m 1 (t3 ) + Llk1 i2pri1 (t3 ) − Lm 2 i2m 2 (t3 ) 2 2 2 1 − Llk2 i2pri2 (t3 ) . (31) 2

where im 1 (t3 ) = 4LVmi n1 T s , ipri1 (t3 ) = im 1 (t3 ), ipri2 (t3 ) =

∞ 4V i n Vo 2 Ts N =1,3,5... N π |Z i n 2 (j N ω S )| sin (θpri2 N ), im 2 (t3 ) = n 2 L m 2 4

443 444

8

446 447 448

449 450

− n 2VLo m2 2 T2s ( π −πθ m 2 ), and θm 2 is the phase angle of the primary voltage vpri2 with respect to that of the voltage vab . The capacitive energy stored in the output capacitors of one pair of primary switches EC can be calculated by 1 EC = 2 × Coss Vin2 . (32) 2 By using (30)–(32), the magnetizing inductance Lm 1 can be calculated by 

4Ts 2 2 2 2 2Coss Vin + Lm 2 im 2 (t3 ) + Llk2 i (t3 ) Lm 1 Vin − Lm 1 − Llk1 ≤ 0.

452 453 454 455 456 457 458 459 460 461 462

Since all of the resonant network parameters for the FBLLC2 and FBLLC1 converters have been decided through step 1 to step 6, the inductive energy EL only depends on the magnetizing inductance Lm 1 . In order to make the turn-off currents of the primary switches as small as possible, the magnetizing inductance Lm 1 needs to be as large as possible in order to achieve a nearly ZCS condition. Therefore, the magnetizing inductance Lm 1 can be calculated from (33) as (34) as shown at the bottom of this page. In order to verify the full soft-switching condition during the CC mode operation, the input impedance Zin (s) of the proposed converter needs to be drawn by using

463 464 465 466

Zin1 (s) Zin2 (s) . Zin1 (s) + Zin2 (s)

The input impedance of the FBLLC1 resonant converter Zin1 (s) and the input impedance of the FBLLC2 resonant converter Zin2 (s) can be calculated by using (36) and (37), respectively Zin1 (s) = sLlk1 +

468 469

470 471 472 473 474

1 sLm 1 Rac1 + sCr 1 sLm 1 + Rac1

1 sLm 2 Rac2 + + sCr 2 sLm 2 + Rac2

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Zin2 (s) = sLlk2 467

(35)

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Zin (s) =

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451

(33)

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(36) (37)

where Rac1 and Rac2 are the reflected load resistance of the FBLLC1 converter and FBLLC2 converter, respectively, and it can be expressed by (38) and (39), respectively Rac1 = n21

8 Vo1 Ro 2 π Vo1 + Vo2

(38)

Rac2 = n22

8 Vo2 Ro . 2 π Vo1 + Vo2

(39)

Fig. 11 shows the input impedance Zin of the proposed converter with varying battery impedance during the CC mode operation drawn with the specification in Table II and the circuit parameters in Table II. As shown in Fig. 11, at the switching frequency, the phase of input impedance Zin always lies in

Fig. 11. Input impedance variation of the proposed converter with varying battery impedance during the CC mode operation. (a) Phase of input impedance. (b) Magnitude of input impedance.

between 3°–8° over the wide range of battery impedance variation. Therefore, the proposed converter can achieve ZVS and nearly ZCS for all of the primary switches during the CC mode operation. V. IMPACT OF THE TOLERANCES IN RESONANT COMPONENTS AND CLOSED-LOOP CONTROL SCHEME OF THE PROPOSED CONVERTER

Lm 1 =

476 477 478

479 480 481

A. Impact of the Tolerances in Resonant Components

482

One important factor needed to be considered in implementing the proposed converter is the tolerances of resonant components, which can affect the output current gain and output voltage gain of the converter. In the design procedure shown in Section IV, the leakage inductance and the magnetizing inductance of the transformer are measured after the transformer is actually implemented. Then, the values of the resonant capacitors (Cr 1 , Cr 2 , and Cr 3 ) are calculated again depending

483

 1+

475

   2 s 1 + Llk1 2Coss Vin2 + Lm 2 i2m 2 (t3 ) + Llk2 i2pri2 (t3 ) 8T Vin    2 s 2 2Coss Vin2 + Lm 2 i2m 2 (t3 ) + Llk2 i2pri2 (t3 ) 4T . Vin

(34)

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VU AND CHOI: NOVEL DUAL FB LLC RESONANT CONVERTER FOR CC AND CV CHARGES OF BATTERIES FOR ELECTRIC VEHICLES (JULY 2017)

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TABLE III VARIATION IN THE RESONANT FREQUENCIES DUE TO THE TOLERANCE OF THE RESONANT CAPACITORS

ro

Fig. 13. Variation in the output voltage gain of the proposed converter according to the variation in the resonant frequencies.

Fig. 12. Change of output current according to the variation of resonant frequencies.

494 495 496 497 498 499 500 501 502 503 504 505 506 507 508 509 510 511 512 513 514 515 516 517 518 519 520 521

EP

493

on the leakage inductances and the magnetizing inductances of the transformers, and the actual resonant capacitors are selected by using the inductance values and the switching frequency as shown in (27)–(29). Therefore, the resonant frequencies fr 1 L L C 1 , fr 2 L L C 2 , and fr 1 L L C 3 are only affected by the tolerance of the resonant capacitors. The following analysis shows as to how much the tolerance of the resonant capacitors affects the variation in the output current and output voltage gains. If the resonant capacitors have ±5% tolerance in theirs values (typical value of J-type), the resulting variation in the resonant frequencies can be calculated as shown in Table III. When the value of resonant frequency varies from 97.6% (48.8 kHz) to 102.6% (51.3 kHz), the output current gain in CC mode operation is varied from 95% (7.4 A) to 105% (8.2 A), as shown in Fig. 12. Similarly, when the value of resonant frequency varies from 97.6% (48.8 kHz) to 102.6% (51.3 kHz), the output voltage gain in CV mode operation is varied from 102% (428.4 V) to 98% (411.6 V), as shown in Fig. 13. As shown in Figs. 12 and 13, the proposed converter still exhibits desired characteristics even though the resonant capacitors have a small deviation in their values. In the CC mode operation, it is important to keep the current constant and the switching frequency needs to be adjusted to find out the exact resonant frequency for the FBLLC2 converter. In this case, the output voltage of the FBLLC1 may not be exactly 210 V but it is not important as long as the output voltage of FBLLC1 is lower than that of the battery. In the CV mode operation, switching frequency needs to be adjusted to keep the output voltage (420 V) constant and the output voltage of each converter may not be exactly 210 V. In this case, a closed-loop control is essential to

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491 492

Fig. 14.

Closed-loop control scheme of the proposed converter.

adjust the switching frequency in order to obtain the suitable current and voltage gains.

523

B. Closed-Loop Control Scheme of the Proposed Converter

525

Once the resonant tanks are designed suitably according to the design procedure presented in the Section IV, the CC and CV mode charge can be simply implemented with a fixed switching frequency and guaranteed the full soft-switching condition. However, it is only possible under the assumption that the input voltage is always constant and there is no tolerance of resonant components. Hence, the operation of the proposed converter deviates from the perfect resonant condition slightly because the input voltage varies in a range of ±5%. A closed-loop control is necessary to implement the CC and CV mode charge operation properly. Therefore, a simple PI control scheme is adopted as shown in Fig. 14. The controller is composed of a PI controller, a mode transfer switch, and a limiter. An anti-wind-up scheme is adopted to reduce the overshoot and transient time caused by the limiter operation.

522

524

526 527 528 529 530 531 532 533 534 535 536 537 538 539 540

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TABLE IV DESIGN PARAMETERS OF THE PROPOSED CONVERTER Parameter

Value

EP

ro

Fig. 15. Bode plots of the proposed converter in CC mode operation under varying load conditions.

50 kHz IPW65R041CFD 1:0.55 13.2 μH 345μH 730 nF PQ50/50 1:0.55 128 μH 272 μH 25 nF 54 nF PQ50/50 HFA50PA60C

of

Switching frequency (fs ) Primary switches (S 1 –S 4 ) Turns ratio of TR 1 (1 : n 1 ) Leakage inductance of TR 1 (L lk 1 ) Magnetizing inductance of TR 1 (L m 1 ) Resonant capacitor of FBLLC1 (C r 1 ) Core size of TR 1 Turn ratio of TR 2 (1 : n 2 ) Leakage inductance of TR 2 (L lk 2 ) Magnetizing inductance of TR 2 (L m 2 ) Resonant capacitor of FBLLC2 (C r 2 ) Resonant capacitor of FBLLC2 (C r 3 ) Core size of TR 2 Rectifier diodes (D 1 –D 8 )

Fig. 17.

Waveforms of the switch S 1 .

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CV mode operation is 0.04 and 12, respectively.

Fig. 16. Bode plots of the proposed converter in CV mode operation under varying load conditions.

541 542 543 544 545 546 547 548 549 550 551 552 553 554 555

When the battery voltage is lower than the maximum charge voltage, 420 V in this research, the current controller is activated to charge the battery with a constant current. Once the voltage of the battery reaches its maximum charge voltage (420 V), the controller automatically switches its mode to the CV mode charge by turning ON the switch S M o de and by activating the voltage controller. In order to design the PI controllers, open-loop transfer functions for CC mode and CV mode are obtained by using the ac sweep function in PSIM software as shown in Figs. 15 and 16, respectively. Based on the obtained bode plots, PI controllers for the CC mode and the CV mode have been designed as shown in (40) and (41), respectively. The Kp value and Ki value of the current controller for CC mode operation is 0.02 and 2, respectively. The Kp value and Ki value of the voltage controller for

2 s 12 Gcv (s) = 0.04 + . s Gci (s) = 0.02 +

556

(40) (41) 557

VI. EXPERIMENTAL RESULTS

558

A 3.3-kW prototype battery charger has been implemented to validate the performance of the proposed dual FBLLC resonant converter. The design parameters are shown in Table IV.

559

561

A. CC Mode Operation

562

Figs. 17–21 shows the key waveforms of the proposed converter at each part during the CC mode operation. (VDC = 400 V, Vo = 250 V, Po = 1.95 kW, Io = 7.8 A, and fs = 50 kHz). Fig. 17 shows the voltage and current waveforms of the primary switch S1 . They clearly show that the ZVS turn-on and

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VU AND CHOI: NOVEL DUAL FB LLC RESONANT CONVERTER FOR CC AND CV CHARGES OF BATTERIES FOR ELECTRIC VEHICLES (JULY 2017)

Primary side waveforms of TR 1 and TR 2 .

of

Fig. 18.

11

Secondary side waveforms of the transformers TR 1 and TR 2 .

EP

Fig. 19.

ro

Fig. 22. Efficiency plots in CC mode operation of the proposed converter and conventional FBLLC resonant converter.

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Fig. 23. Waveforms of the switch S 1 at light load condition (V D C = 400 V, V o = 420 V, P o = 660 W, and fs = 50 kHz).

Fig. 20.

Fig. 21.

Waveforms of the diodes D 1 and D 5 .

Output current and output voltage in CC mode operation.

nearly ZCS turn-off of the primary switches are achieved in the CC charge operation. Primary and secondary side waveforms of the transformer TR1 and TR2 are shown in Figs. 18 and 19, respectively. The rms value of the primary current ipri2 RM S is 5 A as shown in Fig. 18, and the leakage inductance of the transformer TR2 (Llk2 ) is 128 μH as shown in Table IV. It can be seen that the rms value of the primary current ipri2 RM S is successfully limited by the design in step 2 in Section IV. Fig. 20 shows that the rectifier diodes D1 and D5 achieve ZCS turn-off and no reverse recovery. Fig. 21 shows the output current and voltage waveforms of the proposed converter. An efficiency plot of the proposed converter in the CC charge operation is shown in Fig. 22. This plot shows that a maximum efficiency of 98% was achieved. It also shows that the proposed converter has a higher efficiency when compared to the conventional FBLLC converter due its superior soft-switching performance.

569 570 571 572 573 574 575 576 577 578 579 580 581 582 583 584 585 586

B. CV Mode Operation

587

Figs. 23 and 24 show the waveforms of the primary switch S1 in the CV charge operation under light and heavy load

589

588

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ro

Fig. 24. Waveforms of the switch S 1 at heavy load condition (V D C = 400 V, V o = 420 V, P o = 3.3 kW, and fs = 50 kHz).

Output current and output voltage in CV mode operation.

of

Fig. 27.

Primary side waveforms of the transformers TR 1 and TR 2 .

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Fig. 25.

EP

Fig. 28. Efficiency plots in CV charge operation of the proposed converter and conventional FBLLC resonant converter.

Fig. 26.

590 591 592 593 594 595 596 597 598

Waveforms of the diodes D 1 and D 5 .

conditions, respectively. Under both conditions, the switch S1 achieves ZVS turn-on, but the turn-off current is relatively large. Fig. 25 shows waveforms of the primary sides of the transformers in both the FBLLC1 and FBLLC2 converters of the proposed converter during the CV charge operation. It is shown that both the resonant converters operate under perfect resonant conditions. The waveforms of the rectifier diodes D1 and D5 are shown in Fig. 26. Both the rectifier diodes achieve ZCS turn-off. As a result, there is no reverse recovery loss. Fig. 27 shows the

output current and output voltage waveforms of the proposed converter. It should be noted that the loss at the switch S M o de is negligible since there is only a small conduction loss (0.8 W in this case) and no switching loss during the CV mode operation. An efficiency plot of the proposed converter in the CV charge operation is shown in Fig. 28. It shows that a maximum of 97% efficiency was achieved. It also shows that the proposed converter exhibits a slightly higher efficiency when compared to the conventional FBLLC converter. C. Comparison of the Proposed Converter With Other Topologies In order to prove the superiority of the proposed converter, a comparison has been made between the proposed converter and the other conventional LLC resonant converters. Table V provides a comprehensive comparison of the proposed converter with other topologies in relation to the component counts, the complexity of control, and the soft-switching characteristics. A loss comparison between the proposed converter and the conventional FBLLC resonant converter designed with same specification is given in Fig. 29. The losses of the conventional FBLLC converter and the proposed converter are calculated at the end of CC mode charge (VDC = 400 V, Vo = 420 V, Po = 3.3 kW and Io = 7.8 A). As shown in Table V and Fig. 29, the proposed converter has many advantages in terms of losses though the component count of the proposed converter is larger than those of the other topologies.

599 600 601 602 603 604 605 606 607

608 609 610 611 612 613 614 615 616 617 618 619 620 621 622 623 624

VU AND CHOI: NOVEL DUAL FB LLC RESONANT CONVERTER FOR CC AND CV CHARGES OF BATTERIES FOR ELECTRIC VEHICLES (JULY 2017)

TABLE V COMPARISON OF THE PROPOSED CONVERTER WITH OTHER TOPOLOGIES Conventional LLC [1], [2]

References [20], [21] Fixed

Fixed

Wide range

Number of additional switches

1

0

2

Switching loss of additional switches Control scheme of additional switches Number of the resonant tanks

None

None

High

Simple

None

Complex

2

1

1

Number of the transformers Switching-on of the primary switches in CC mode Switching-off of the primary switches in CC mode Circulating current in CC mode

2

1

1

ZVS

ZVS

ZVS

Nearly ZCS High turn-off High turn-off current current Small Large Medium

[1] R. Beiranvand, B. Rashidian, M. R. Zolghadri, and S. M. H. Alavi, “Using LLC resonant converter for designing wide-range voltage source,” IEEE Trans. Ind. Electron., vol. 58, no. 5, pp. 1746–1756, May 2011, doi: 10.1109/TIE.2010.2052537. [2] F. Musavi, M. Craciun, D. S. Gautam, W. Eberle, and W. G. Dunford, “An LLC resonant DC-DC converter for wide output voltage range battery charging applications,” IEEE Trans. Power Electron., vol. 28, no. 12, pp. 5437–5445, Dec. 2013, doi: 10.1109/TPEL.2013.2241792. [3] R. Beiranvand, B. Rashidian, M. R. Zolghadri, and S. M. H. Alavi, “A design procedure for optimizing the LLC resonant converter as a wide output range voltage source,” IEEE Trans. Power Electron., vol. 27, no. 8, pp. 3749–3763, Aug. 2012, doi: 10.1109/TPEL.2012.2187801. [4] J. Deng, S. Li, C. C. Mi, and R. Ma, “Design methodology of LLC resonant converter for electric vehicle battery chargers,” IEEE Trans. Veh. Technol., vol. 63, no. 4, pp. 1581–1592, May 2014, doi: 10.1109/TVT.2013.2287379. [5] H. P. Park and J. H. Jung, “Modeling and feedback control of LLC resonant converter at high switching frequency,” J. Power Electron., vol. 16, no. 3, pp. 849–860, 2016. [6] B. Yang, F. C. Lee, A. J. Zhang, and G. Huang, “LLC resonant converter for front end DC/DC conversion,” in Proc. IEEE 17th Annu. IEEE Appl. Power Electron. Conf. Expo., 2002, vol. 2, pp. 1108–1112, doi: 10.1109/APEC.2002.989382. [7] C. W. Tsang, M. P. Foster, D. A. Stone, and D. T. Gladwin, “Analysis and design of LLC resonant converters with capacitor-diode clamp current limiting,” IEEE Trans. Power Electron., vol. 30, no. 3, pp. 1345–1355, Mar. 2015, doi: 10.1109/TPEL.2014.2317653. [8] K. H. Yi, and G. W. Moon, “Novel two-phase interleaved LLC seriesresonant converter using a phase of the resonant capacitor,” IEEE Trans. Ind. Electron., vol. 56, no. 5, pp. 1815–1819, May 2009, doi: 10.1109/TIE.2008.2011310. [9] Z. Hu, Y. Qiu, Y. F. Liu, and P. C. Sen, “A control strategy and design method for interleaved LLC converters operating at variable switching frequency,” IEEE Trans. Power Electron., vol. 29, no. 8, pp. 4426–4437, Aug. 2014, doi: 10.1109/TPEL.2014.2300165. [10] H. Wang, S. Dusmez, and A. Khaligh, “Maximum efficiency point tracking technique for LLC-based PEV chargers through variable DC link control,” IEEE Trans. Ind. Electron., vol. 61, no. 11, pp. 6041–6049, Nov. 2014, doi: 10.1109/TIE.2014.2311399. [11] W. Feng, F. C. Lee, and P. Mattavelli, “Simplified optimal trajectory control (SOTC) for LLC resonant converters,” IEEE Trans. Power Electron., vol. 28, no. 5, pp. 2415–2426, May 2013, doi: 10.1109/ TPEL.2012.2212213. [12] R. Beiranvand, B. Rashidian, M. R. Zolghadri, and S. M. H. Alavi, “Optimizing the normalized dead-time and maximum switching frequency of a wide-adjustable-range LLC resonant converter,” IEEE Trans. Power Electron., vol. 26, no. 2, pp. 462–472, Feb. 2011, doi: 10.1109/TPEL.2010.2068563. [13] Z. Fang, T. Cai, S. Duan, and C. Chen, “Optimal design methodology for resonant converter in battery charging applications based on time-weighted average efficiency,” IEEE Trans. Power Electron., vol. 30, no. 10, pp. 5469–5483, Oct. 2015, doi: 10.1109/TPEL.2014.2379278. [14] Z. G. Liang, R. Guo, G. Y. Wang, and A. Huang, “A new wide input range high efficiency photovoltaic inverter,” in Proc. IEEE Energy Convers. Congr. Expo., 2010, pp. 2937–2943, doi: 10.1109/ECCE.2010.5618217. [15] J. H. Kim, C. E. Kim, J. K. Kim, J. B. Lee, and G. W. Moon, “Analysis on load-adaptive phase-shift control for high efficiency full-bridge resonant converter under light-load conditions,” IEEE Trans. Power Electron., vol. 31, no. 7, pp. 4942–4955, Jul. 2016, doi: 10.1109/TPEL.2015. 2462077. [16] B. McDonald and F. Wang, “LLC performance enhancements with frequency and phase shift modulation control,” in Proc. IEEE Appl. Power Electron. Conf. Expo., Mar. 2014, pp. 2036–2040., doi: 10.1109/ APEC.2014.6803586. [17] Y. K. Lo, C. Y. Lin, M. T. Hsieh, and C. Y. Lin, “Phase-shifted full-bridge series-resonant DC-DC converters for wide load variations,” EEE Trans. Ind. Electron., vol. 58, no. 6, pp. 2572–2575, Jun. 2011, doi: 10.1109/TIE.2010.2058076. [18] H. Wu, T. Mu, X. Gao, and Y. Xing, “A secondary-side phaseshift-controlled resonant converter with reduced conduction loss at normal operation for hold-up time compensation application,” IEEE Trans. Power Electron., vol. 30, no. 10, pp. 5352–5357, Oct. 2015, doi: 10.1109/TPEL.2015.2418786.

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Reverse recovery loss of rectifier None Exist None diodes in CC mode Switching-on of the primary ZVS ZVS ZVS switches in CV mode Switching-off of the primary High turn-off High turn-off High turn-off switches in CV mode current current current Circulating current in CV mode Large Large Medium

REFERENCES

of

Switching frequency

Proposed

625 626 627 628 629 630 631 632 633 634 635 636 637 638 639 640

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Fig. 29. Loss comparison between the proposed converter and conventional FBLLC converter at full load condition.

VII. CONCLUSION

This paper introduced a novel dual FBLLC resonant converter suitable for implementing the CC and CV charge of the batteries. By designing the two resonant converters to resonate at lower and higher resonant frequencies, respectively, it is possible to obtain the advantages of each operation, which results in ZVS and nearly ZCS of the primary switches in the CC charge operation. The CV charge is implemented by operating both the converters at the higher resonant frequency by transforming the resonant tank of the FBLLC2 resonant converter. The converter operates with no switching losses at the primary switches and no switching frequency variation over the entire charge process. Its high efficiency over a wide range of output power makes it suitable for battery charge applications, such as electric vehicles and energy storage systems.

13

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Hai-Nam Vu received the B.S. degree from the Hanoi University of Science and Technology, Hanoi, Vietnam, in 2014, and the M.S. degree from Soongsil University, Seoul, South Korea, in 2017, both in electrical engineering. From 2014 to 2015, he was an Assistant Researcher in the Centre for Technology Innovation, Hanoi University of Science and Technology. His research interests include dc–dc converters dealing with the renewable sources and batteries for energy storage system or hybrid electric vehicles, soft-switching techniques for the pulse width modulation/pulse-frequency modulation converters, and resonant converters.

Woojin Choi (S’00–M’05) was born in Seoul, South Korea, in 1967. He received the B.S. and M.S. degrees from Soongsil University, Seoul, South Korea, in 1990 and 1995, respectively, and the Ph.D. degree from Texas A&M University, College Station, TX, USA, in 2004, all in electrical engineering. From 1995 to 1998, he was a Research Engineer in Daewoo Heavy Industries, Seoul, South Korea. In 2005, he joined the School of Electrical Engineering, Soongsil University, Seoul, South Korea. His research interests include modeling and control of electrochemical energy sources, such as fuel cells, batteries, and supercapacitors; power conditioning technologies in renewable energy systems; and dc–dc converters for electric vehicles and fuel cells. Dr. Choi is an Associate Editor of the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS and a Publication Editor of the Journal of Power Electronics of the Korean Institute of Power Electronics.

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[19] K. Zheng, D. Zhou, J. Li, L. Li, and Y. Zhao, “A digital self-sustained phase shift modulation control strategy for full-bridge LLC resonant converters,” J. Power Electron., vol. 16, no. 3, pp. 915–924, 2016. [20] X. Sun, X. Li, Y. Shen, B. Wang, and X. Guo, “A modified high-efficiency LLC converter with two transformers for wide input-voltage range application,” IEEE Trans. Power Electron., vol. 28, no. 4, pp. 1946–1960, 2016, doi: 10.1109/TPEL.2012.2201959. [21] H. Huau, X. Fang, F. Chen, Z. J. Shen, and I. Batarseh, “A dual-bridge LLC resonant converter with fixed-frequency PWM control for wide input applications,” IEEE Trans. Power Electron., vol. 28, no. 1, pp. 1946–1960, Apr. 2013, doi: 10.1109/TPEL.2016.2530748. [22] J. Dodge, “Power MOSFET tutorial,” Adv. Power Technol., Beach, FL, USA, Appl. Note APT0103.

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715 716 717 718 719 720 721 722 723 724 725 726 727

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Queries

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Q1. Author: Please confirm if “(July 2017)” should be present in the article title. Q2. Author: Please provide the expanded form of “CC,” “CV,” “ZCS,” and “ZVS” at their first occurrence in the text. Q3. Author: Please provide the pincode of “Hanoi University of Science and Technology” in the affiliation of H.-N. Vu. Also, Please check if the affiliations are okay as set for correctness. Q4. Author: Please check if (19) should be replaced with (20) in the sentence starting rom “Since the FBLLC2 resonant converter operates . . . .” Q5. Author: The captions of Figs. 20 and 26 are similar. Please check. Q6. Author: Please provide year in Ref. [22].

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