Advanced SPICE Modeling of Large Power IGBT Modules R. Azar, F.Udrea, M.De Silva, G.Amaratunga
W.T. Ng, F. Dawson
W. Findlay, P. Waind
Cambridge University Department of Engineering University of Toronto, Department of Trumpington Street, Cambridge, UK CB2 1PZ Electrical and Computer Engineering, Tel. +44-1223-332672, Fax. +44-1223-332662 Toronto, ON, Canada M5S 3G4 Tel. (416) 978-6249 Fax. (416) 971-2286
[email protected]
Dynex Semiconductors Lincoln, UK Tel. +44-1522-502760 Fax. +44-1522-502747
Abstract— An enhanced IGBT model based on the Kraus model with new derivations based on an extra parameter accounting for p-i-n injection was developed to allow simulation of both Trench and DMOS IGBT structures. Temperature dependence was also implemented in the model. The model was validated against steady state and transient measurements done on an 800A, 1.7kV Dynex IGBT module at 25°°C and 125°°C. The Spice model has also shown excellent agreement with mixed mode MEDICI simulations. The Spice model takes also into account for the first time the parasitic thyristor effect allowing the DC and dynamic temperature dependent latchup modeling of power modules as well as their temperature dependent Safe Operating Area. Keywords—IGBT model; PSPICE; Trench IGBT Simulation
Fig 1. The original Kraus Model and Charge Subcircuit
I. INTRODUCTION The industry has until now favored the Kraus IGBT model for use with SPICE and Saber because it offers an excellent trade-off between speed and complexity in dc and transient simulations. However, it does not account for temperature dependence or self-heating effects and does not include any temperature dependent SOA assessments. It is also limited to DMOS type IGBTs. There is great industry demand for simulations of large area IGBTs, and in particular Trench IGBTs, but there are few or no available fast and robust SPICE or Saber models for these structures at the present time. An enhanced IGBT model based on the Kraus model [1] with new derivations accounting for p-i-n injection was developed for the simulation of both DMOS and Trench structures. Results were validated against measurements on 1.7kV Dynex IGBT modules and MEDICI simulation. The parasitic thyristor effect has also been modeled along with temperature dependence, thereby allowing temperature dependent SOA simulations in SPICE. II.
CHARGE EQUATIONS
A. The original assumption in the Kraus Model Fig. 1 shows the basic IGBT Kraus model and base charge subcircuit [1]. The collected hole current Ipc, the emitter electron current Ine and the variable base resistance Rb are all controlled by the base charge Qb. The current Inc depends on the MOSFET threshold voltage and gain factor which in turn reflect the layout and gate density of the IGBT.
The reverse saturation current Ise controls the injection efficiency and carrier concentration at the anode end of the base. At the cathode end of the base, the model inherently assumes a zero carrier concentration in the on state. B. Accounting for p-i-n injection It has been shown in [2] that p-i-n injection has a significant contribution to the current in DMOS structures and is also largely responsible for the lower on-resistance in Trench structures due to their large accumulation area. Accounting for p-i-n injection will require making changes in the base charge distribution equation while preserving model simplicity and robustness. Solving the ambipolar diffusion equation for a non-zero carrier concentration at the cathode end and integrating over the base width W will yield the solution for the steady-state base charge Qb0:
Qb 0 = qAni L( po + pw ) tanh
W 2L
(1)
Where L is the ambipolar diffusion length and pw and po are the carrier concentrations at the cathode and anode edges of the base respectively. Under high injection conditions, the charge neutrality condition at the emmitter-base junction is: n0 = p0 = ni e
qV
2 kT
(2)
where V is the voltage dropped through the junction. Substituting this result into the diode current equation p0 can
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be derived as a function of Ine and Ise. Using a similar approach for the n+accumulation-n-base junction will also yield a solution for pw:
po = ni
I ne I se
pw = ni
I nc I sc
(3)
where Isc is the reverse saturation current of the p-i-n diode. Considering that in reverse bias Isc is equal to the reverse current through the p+emmitter-n-base junction, it will be assumed that Isc=Ise. This will tend to overestimate Isc since the total area of the emmitter-base junction is larger than the area of the p-i-n diode. Furthermore, as Inc includes current contributions from both the MOSFET channel and the p-i-n effect, using Inc in (3) will tend to overestimate pw. This overestimation can be corrected by introducing a scaling factor “γ” for pw, as follows: γ =
Ap
characteristics of the IGBT under the new charge conditions derived in (6). Note that in more complex models such as the PIN-PNPMOS model [2], more accurate expressions are derived for pw. However since the approximation proposed here allows immediate insertion within the Kraus model, the resulting model benefits from the Kraus model’s robustness and stability in circuit simulators such as PSPICE or SABER while requiring minimum extra computational power. C. Simulation Results A comparison of the on-state currents of a DMOS IGBT using the original Kraus model and a Trench IGBT using the charge equations proposed here is shown in Fig. 2. The substantial drop in on-resistance with increasing γ is consistent with the MEDICI results obtained in [2].
(4)
Acell
where Ap is the accumulation layer area and Acell is the total cell area. This factor also introduces a dependence of the charge on the design aspect ratio and technology used (DMOS or Trench). Substituting this new boundary condition in (1) and solving for Ine will yield the following result:
Qb 0 − γ I nc I ne = I se qAni L tanh W 2L
(
2
(5)
)
Introducing this change into the Kraus model’s recombination equation [1] and solving for the charge will yield the final solution for the base charge Qb0:
2Qbd − Qb0 =
η2 2α2Td
F3 −η + F32 − 2ηF3 + 3Td Ise
(6)
Qbd
[qAnL tanh(W 2L)]
2
i
Fig. 2. Drain transfer characteristic for VG = 15V showing the original DMOS Kraus model with γ = 0 and the proposed model adapted for Trench IGBTs with γ = 0.8.
III. TEMPERATURE DEPENDENCE The temperature dependence of the injection efficiency, represented by the current Ise in the Kraus model (Fig 1.), was derived from the ideal diode equation. The temperature dependence of the diffusion length, lifetime, mobility and intrinsic carrier concentration of silicon has been documented in [5]. The result shows the product of a linear and an exponential temperature component as follows:
I se = Aq
Where:
η = 2αTdγ Inc
α2 =
Ise
[qAnLtanh(W 2L)]
AT 2.5 = (3.88 ⋅10 ) ⋅ 0 ND
2
16 2
i
F3 = 1 +
Td
τb
D p ni2 τ p Nd
Qbd = Qb + Td Inc
Note that if γ=0 then (6) reduces to the standard Kraus model charge expression of (1). Substituting this new charge derivation in the current equations of the original PSPICE Kraus model allows one to simulate the forward and transient
kqµ no
τo
⋅T ⋅e
−1400 T
(7)
where To is the room temperature and µn0 and τ0 are the room temperature electron mobility and base lifetime respectively. This reflects an increase in Ise with increasing temperature. This now allows one to simulate the IGBT at any given temperature. A comparison of this new model and the mixed mode MEDICI model to actual on-state and transient measurements done on an 800A, 1.7kV Dynex IGBT module at 25°C and 125°C shows very good agreement (Figs 3 and 4).
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I (A), V/10 (V), E (mJ)
The turn off energy predicted by the SPICE model is also in excellent agreement with the energy predicted by MEDICI and the actual measurements. Adding the thermal model of the package in the form of an RC circuit allows one to account for the self-heating effect during DC and transient operation. Vg=20V Ianode (A)
Vg=15V
Emedici=29mJ Edynex=32mJ EPSpice=30mJ
Vg=12 DynexMeasurements
Vg=10V (b)
Vanode (V)
Fig. 4.Measured turn-off characteristic and energy of Dynex 1.7kV modules v.s. mixed mode model and proposed SPICE model at (a) 25°C and (b) 125°C.
Medici Model
(a)
Ianode (A)
Vg=20 V
Vg=15V Vg=12V Vg=10V
Vanode (V)
Medici Model
(b) Fig. 3. Drain transfer characteristic of the new model v.s. the MEDICI results and actual measurements done on a 1.7kV IGBT module at (a) 25°C and (b) 125°C.
IV. LATCHUP MODELING Adding the modeling of the latch up effect is essential in order to evaluate a product for practical applications. Modifying the Kraus model to include the latchup behavior implies adding the parasitic NPN bipolar transistor to the basic circuit with the emitter connecting to the source of the MOSFET and separated from the base by the source resistance while the collector connects directly to the base. A. Steady-State Latchup The solution proposed here consists in a change of the cathode position, so that it contacts the source of the MOSFET directly, as shown in Fig. 5. The source resistance Rs, originally present in the model to account for imperfections in the MOSFET channel, is now used as the latchup p-well resistance, thereby isolating Ipc, the hole current responsible for latchup. Since Ipc is controlled by a current source the prelatch up characteristics will remain unchanged. After NPN turn-on the thyristor action is modeled by measuring the NPN current and feeding it to the charge accumulation subcircuit. Anode P+ Ise
I (A), V/10 (V), E (mJ)
N+
Emedici=24mJ EPspice=25mJ Edynex=27mJ
Cox
Ved
Edep
NPN
Inpn
Gate
Ib=Ved/Rb -
Dynex Measurements
-
New Cathode (a)
(a)
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Vbe
Nbase Ipc Pwell
Rs
Cgs
V(b)
+
1µF
Ine
Inc
Vanode (V), Ipc(A), Inpn (A), Itot (A)
Base node
V(b)=Q(b)/1u
Inpn
τb/1u
(b) Fig. 5.(a) Kraus model circuit with added npn BJT and new cathode position, (b) base charge sub-circuit with the added latchup current component.
B. Dynamic Latchup In the case of transient operation, two main current components contribute to an eventual latch up: •
the increase of Ipc resulting from an increased αpnp when the gate current Inc is turned off
•
the displacement current required to deplete the pwellnbase junction during turn-off
As can be seen from Fig. 5, the modification in the cathode position also allows both these two contributions to go through Rs during a transient period. C. Latchup Simulation Results Extracting the value of Rs can be done in MEDICI by plotting both the hole quasi-fermi potential and the current Ipc along the pwell depletion edge. Rs can also be approximated by integrating the silicon resistivity along the current path in the undepleted region of the pwell. The value of the vbe turn on voltage can be calculated from the junction doping levels. Note that since the actual Rs found for the Dynex module is less than l50µΩ, no latchup curves could be simulated or measured. To demonstrate the latchup features of the model in SPICE simulations, a higher, hypothetical value of Rs=10.8mΩ was used. Results for static and dynamic latchup are shown in Fig. 6. The latchup current threshold is found to be constant irrespective of Vg in Fig. 6(a). Latchup at lower gate voltages due to increasing temperature can be simulated by adding the package thermal circuit in SPICE. In Fig. 6(b), since the forward voltage drop decreases below the on-state value as a result of the thyristor effect, the total current increases slightly. Overall this allows one to analyze and extract the parasitic components of large power IGBT modules to predict accurately their switching behavior.
Ianode (A)
NPN turn-on
Voltage drop due to thyristor action
Fig. 6. (a) DC latch up curves for hypothetical Rs = 10.58mΩ, (b) Dynamic latchup showing increase in post-latchup current due to snap back behaviour.
V. CONCLUSION A new IGBT model was proposed. An extra parameter γ representing the ratio of the accumulation layer area to the active region area was introduced to account for the IGBT aspect ratio and p-i-n effect. The resulting model can accommodate Trench structures and allow higher accuracy on DMOS structures. Temperature dependence was introduced in the model. The model was also adapted to include the parasitic thyristor effect, allowing accurate simulations of forwardbiased, reverse biased and short-circuit SOAs. Overall this new model not only benefits from the simplicity, speed and robustness of the Kraus model, but also incorporates temperature change effects and latch up modeling, a necessary feature for the industrial benchmarking of IGBT modules. Furthermore, it answers the industry’s demand for an easy way to evaluate and/or simulate the on state, transient, and SOA curves of large power modules such as IGBTs and in particular Trench IGBTs. ACKNOWLEDGMENTS R. Azar acknowledges the award of a full PhD Scholarship from Dynex Semiconductors, UK. F. Udrea acknowledges the award of an EPSRC Advanced Fellowship (AF/100027). W.T. Ng acknowledges CITO and the Ministry of Energy, Science and Technology of Ontario. F. Dawson and W. T. Ng acknowledge the financial support given by Dynex Semiconductor, UK. The authors also acknowledge TMA for the use of the MEDICI software. REFERENCES [1]
Vg=12V Vg=15V
[2]
[3]
Vg=10V [4] Vanode(V)
(a)
Ipc increases as Vg
