An AC-DC Multilevel Converter Feasible to Traction Application Dalton Honório, Demercil Oliveira, Luiz Henrique Barreto FEDERAL UNIVERSITY OF CEARÁ Centro de Tecnologia - Campus do Pici, Bl. 705 Fortaleza-CE, Brazil Tel.: +55 / (85) – 3366-9581. Fax: +55 / (85) – 3366-9574. E-Mail:
[email protected] URL: http://www.dee.ufc.br
Acknowledgements The authors acknowledge the Brazilian National Council for Research and Development (CNPq) and Coordination of Improvement of Higher Education Personnel (CAPES) for the financial support to this work, besides the Federal University of Ceará for the structural support.
Keywords «Solid State Transformers», «Multilevel Converters», «Traction applications», «AC-DC Converter».
Abstract This paper proposes an ac-dc bidirectional power converter topology consisting of an arrangement of full-bridge converters based on the three-state switching cell using a medium-frequency isolation transformer. The topology description and simulation results for the dynamic conditions are presented and discussed on detail.
Introduction In the past few years, solid state transformers (SST) have become a real possible solution for many applications in power systems. This could be evidenced in traction applications; since this field is recognized as one of the primary topics that deserved significant research effort in towards the replacement of the bulky classical power line frequency transformer (LFT). The main concept involving this approach is to replace the conventional transformer operating at the grid frequency, which is usually lower than 100 Hz, for a topology using a medium-frequency transformer (MFT) designed for an operating frequency around hundreds of hertz. Besides, the structure is supposed to aggregate high input voltage capability.
Fig. 1: Power electronic converters used in traction applications: (a) Classical approach with LFT, (b) SST approach with MFT.
Fig. 1 (a) shows the classical architecture applied in traction systems. This approach requests a bulky transformer that must provide insulation and voltage adaptation for the locomotive machine driver systems. As can be seen, the LFT steps down the input voltage level for one feasible to feed conventional drive converters of the locomotive. On the other hand, Fig. 1 (b) presents an example of the SST technology applied to traction structures. This configuration consists in a three-stage conversion topology with low-voltage dc-link capacitors. The first stage can be decomposed in a series arrangement, in order to employ low dc voltage units, thus reducing the involved cost. In the intermediate dc-dc stage, the transformer operating frequency is higher, as its weight and volume is reduced. Finally, the third stage allows the implementation of traditional machine drive solutions, as in the classical approach shown in Fig. 1 (a). Multilevel topologies are widely used in SST solutions due to the possibility to apply lower voltage capability semiconductors, also allowing the use of identical modules aggregating redundancy and reliability, especially in traction applications. There are three SST types in terms of the number of conversion stages. The first type uses topologies in a single-stage approach. They are usually designed with large number of semiconductors, which typically limits efficiency. Besides, the regulation of the converter variables must to be performed with reduced degree of freedom, thus increasing complexity [1-4]. A first approach was proposed by Mc Murray [5], where bidirectional switches are used with a singlephase tapped transformer. The arrangement is described so that the output frequency must be the same as the input one, thus limiting the application of the converter. Analogously, Enjeti [6] has presented a topology with the same number of switches in a H-Bridge configuration for both sides of a MFT and Drabek [7] has used them in a multiple primary winding transformer. The topologies presented in [8] are also analyzed Ostlund [9], thus resulting in the reduction of 40% in the total weight of the transformer if compared with a low-frequency counterpart. In other solutions, e.g. three-stages [10-22], control decoupling is possible, allowing the individual implementation of the control loops in each stage for a proper functionality. For two-stage solutions [23-28], such decoupling is not always possible to obtain in practice. Therefore, such approach becomes a more complex from the control system point of view if compared with three-stage ones. In fact, the choice of a given SST solution depends on the given application. For instance, Rufer [29] uses cascaded H-bridge (CHB) converters with the aim to obtain split dc links that are connected to dual active bridge (DAB) converters. The outputs are then connected in parallel, resulting in a single dc output voltage across. Other proposal uses the so-called Modular Multilevel Line Converters (M2LC) concept presented by Maquart [30-32], which consists in equal modules of cascaded full-bridge converters associated in a H-bridge structure (each leg are composed by two modules). It is possible to obtain a voltage waveform with low harmonic contents regarding the MFT. However it is necessary high component count depending on the rated specifications. In this context, this paper presents a single-stage ac-dc bidirectional topology converter that can be applied to SSTs focusing on traction applications. The whole topology is a two-stage approach with an ac-dc stage considering it supplies an ac machine drive. The description regarding the first stage of the topology is presented, as simulations results are presented and discussed to validate the assumptions.
Proposed topology The proposed topology is presented in Fig. 2. Fig. 2 (a) shows the basic module of the structure represented by i.
Fig. 2: Proposed topology: (a) Basic arrangement of the proposed topology; (b) Variation of the proposed topology with six switches in the primary side; (c) Generalized approach. The converter can be seen as the result of a combination between the DAB topology [33] and the three state switching cell (3SSC) [34] concept in order to obtain an arrangement that provides good distribution of losses among the semiconductors and minimization of harmonic contents for both voltages and currents. Each module has two parts called submodules, which are represented by SMpri_i in the primary side of the MFT and SMsec_i in the secondary side. Fig. 2 (c) presents the generalized topology for a generic number of modules n. The primary side of the MFT uses series-connected submodules SMpri_i, while the secondary side uses parallel configuration. It is also possible to derive another topology by replacing capacitors C+i and C-i for switches. The resulting structure is presented in Fig. 2 (b). In this case the static converter gain and the total number of modules is reduced to a half. In fact, the topology presented in Fig. (2) (c) has almost 40% less switches for n=6 modules if compared with the one shown in Fig. (2) (a).
Fig. 3: Possible switching states of the proposed topology
In the full-bridge converter, switches (Si1, Si2) and (Si3, Si4) operate complementarily. Therefore, there are four possible switching states in the primary side of the MFT, which are presented in Fig. 3, considering the secondary of the DAB converter as a three level ac voltage source. The switching states are summarized in Table I, as well as the charging condition for capacitors C+i and C-i. From Fig. 3, it can be seen that the multilevel voltage Vmult_i can be modified without changing the voltages across x and y equal to Vpri_i, which is possible by changing the switching vectors between (1,0,1,0) and (0,1,0,1). Analogously, it is possible to modify Vpri_i without changing Vmult_i, with states (1,0,0,1) and (0,1,1,0). Similar behavior is noticed for capacitors C+i and C-i, while it is possible to charge one as the other is discharging. Otherwise, it is possible to charge a single capacitor without modifying the charging state of the remaining one.
Table I: Analysis of the switching states for the proposed converter Iin
Positive semi-cycle
Negative semi-cycle
Si1
Si2
Si3
Si4
Vpri_i
Vmult_i
1
0
1
0
0
1
0
0
1
+ Vdc
0
↑
↓
0
1
1
0
- Vdc
0
↑
↓
0
1
0
1
0
-
Vdc
1
0
1
0
0
-
Vdc
1
0
0
1
- Vdc
0
↓
↑
0
1
1
0
+ Vdc
0
↓
↑
0
1
0
1
0
+
+
Vdc
Vdc
2
VC+i ↑
↓
2 2
2
VC-i
↓
↑
The aforementioned relationships are important when a voltage balancing scheme is necessary for capacitors C+i and C-i, as well as in the individual control of the magnetizing current through each module.
Modulation technique and control strategy Analogously to De Donker [34], power flow between the primary and secondary sides can be controlled by the phase-shift angle Φi between two bridges of each module, as shown in Fig. 2 (a). In addition, PFC is performed with the duty cycle variation in SMpri_i. The technique used in this work is called phase-disposition (PD) modulation [35] (Fig. 4), which provides equal power sharing among all modules and ensures the medium-frequency operation of the transformers. Fig. 4 (a) shows the application of this technique to a single module of the proposed converter. Fig. 4 (b) presents the multilevel waveforms obtained for n={1,2,3}, respectively. It can be noticed that each module contributes with two possible levels besides zero, for the resulting multilevel waveform. Then, the relationship between the number of modules and the maximum number of levels is given by (1). nlevels = 2 n + 1
(1)
Fig. 4: Waveforms used in the converter analysis: (a) adopted modulation technique; (b) multilevel waveform in function of the number of modules. The control strategy used in the proposed converter can be seen in Fig. 5. It aggregates six different aspects: dc link voltage control; PFC control i.e. input current shaping; balance of the dc link voltage; output voltage control; magnetizing current control through the MFT; and balance between voltages VC+i and VC-i.
Fig. 5: Proposed control strategy. Each module has a small value of Φ’i, which is added to the main angle Φ in order to adjust the respective power flow and regulate the dc link voltages with consequent balancing. The modulating signal M is obtained from the external Dc link voltage loop and the internal input current loop added to the parallel middle-point voltage loop.
Simulations results Simulation results are presented considering a small-scale model for future laboratory implementation with six modules (n=6), assuming a switching frequency of 20 kHz, rms ac input voltage of 760 V/60 Hz, dc link voltage across each submodule equal to 400 V, dc output voltage equal to 400 V, and output power of 6 kW. First, the dynamic simulations results are presented considering an inversion of the power flow direction from -100% to 100% of the rated power. Fig. 6 corresponds to the voltages across capacitors C+i and C-i, the output voltage, the input current, as well as the output current. Fig. 6 (a) shows the behavior the voltages across C+i and C-i in the six-module of the converter. A maximum overshoot about 20% with settling time about 300 ms exists. These results evidence the accurate operation of the dc link voltage control loop, once that at the moment of the power flow inversion the voltages tend to follow the reference value while the proper balance of the capacitor voltages in each module is obtained.
Fig. 6: Results for power flow inversion. Similar behavior occurs in Fig. 6 (b) for the output voltage waveform, which is maintained around 400 V. At the moment of the step, there is a voltage ripple equal to about 4 V, which represents 1 % of the reference value. This result validates the operation of the output voltage control loop. Fig. 6 (c) and Fig. 6 (d) presents the input and output current waveforms, respectively. The results show clearly the complete inversion of the power flow in the converter. In fact, the output current changes its phase instantly at 0.61 s, while the input current takes almost three cycles to be stabilized. Simulations results for a 50% load step are shown in Fig. 7. The first result (Figure 7 (a)) shows the voltages across capacitors C+i and C-i, which are maintained constant at 200 V with a ripple of 5 V before the step. The load step effectively occurs at 0.52 s and the voltages present overshoot of about 15% and settling time of 120 ms, with the ripple is about 20 V.
Fig. 7: Results for a 50 % load step.
Fig. 7 (b) presents the output voltage behavior, where it possible to see that output voltage control loop operates properly due to the accurate reference tracking at 400 V, with settling time of 0.8 ms and overshoot lower than 1 %. The behavior of the currents through the output and input sides can be seen in Fig. 7 (c) and Fig. 7 (d), respectively. Their magnitudes vary due to the load step at 0.52 s. In fact, the input current increases in order to supply additional power to the load. The next result has the goal to verify the disturbance rejection of the dc link voltage balancing control loop, which maintains the voltages across C+i and C-i nearly equal. Thus, such loop is able to regulate the average dc link voltage more effectively. The Fig. 8 (a) presents the individual phase shift angles Φi of each module. Initially, all the modules are phase-shifted by -30º. Then, a step is applied at 0.22 s instant considering that both power flow occurs in directions, thus simulating a disturbance in the ideal steady-state operation and resulting in unbalanced voltages across the dc link capacitors as show in Fig. 8 (b). Then, the control loop operates in order to mitigate such disturbances and the correct tracking of the reference value is evidenced after 0.25 s as the same initial steady-state condition is achieved. Such dynamic behavior validates the proper operation of the dc link voltage balancing control loop.
Fig. 8: Results for disturbance rejection of the dc link voltage balancing control loop.
Conclusion This work has presented a new topology for a single-stage bidirectional ca-cc multilevel converter feasible for SST in traction applications. Besides, a possible variation of the proposed topology is shown, as the switches in the primary side of the MFT are replaced by capacitors, consequently reducing overall component count. The simulations results regarding the converter dynamics have evidenced the accurate operation of the control loops and the possible functionalities that could be aggregated in a SST based on this study.
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