Cad and optimization of compact ortho-mode transducers - IEEE Xplore

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 47, NO. 12, DECEMBER 1999

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CAD and Optimization of Compact Ortho-Mode Transducers Michele Ludovico, Bruno Piovano, Associate Member, IEEE, Giorgio Bertin, Member, IEEE, Giovanna Zarba, Member, IEEE, Luciano Accatino, Member, IEEE, and Mauro Mongiardo, Member, IEEE

Abstract— We describe a hybrid computer-aided-design technique, which employs the standard generalized scattering matrix description and a new one, based on the three-dimensional generalized admittance matrix representation, for the efficient and reliable analysis of compact ortho-mode transducers (OMT). A rigorous investigation of the numerical convergence properties of the electromagnetic simulators is carried out by using as a benchmark the measurements of a specifically built structure. The electromagnetic simulators are used for the efficient computer-aided optimization of the OMT and a procedure for the dynamical optimization of such components is introduced and tested by designing several OMT’s operating at L, , and frequency bands. The component design, entirely carried out at computer level, has demonstrated significant advantages in terms of development times and no need of post-manufacturing adjustments. The very satisfactory agreement between experimental and theoretical results further confirms the validity of the proposed technique.

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Index Terms— Antenna accessories, convergence of numerical methods, design automation, mode-matching methods, optimization methods, waveguide components.

I. INTRODUCTION

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RTHO-MODE transducers (OMT’s) represent a key element in the realization of dual-polarization antennas, which are currently employed in order to increase the satellite traffic capacity by simultaneously using two different polarizations (in our case, vertical and horizontal). From the block diagram shown in Fig. 1, it is apparent that the OMT has a significant impact on the entire system performance. Since it generally operates behind a circular horn, its output needs to support dual linear polarizations in a circular waveguide, whereas its two inputs are provided by the feeding rectangular waveguides. As noted in [1, p. 377], despite the common application of dual polarized transmission in high capacity communication systems for several decades, the design of the key component [i.e., the OMT] is treated very unsatisfactorily in the literature. This holds true either for classical microwave books

Manuscript received March 26, 1999; revised July 14, 1999. M. Ludovico, B. Piovano, G. Bertin, G. Zarba, and L. Accatino are with the Microwave Department, Centro Studi e Laboratori Telecomunicazioni S.p.A., 274-10148 Turin, Italy (e-mail: [email protected]). M. Mongiardo is with the Istituto di Elettronica, I-06100 Perugia, Italy (email: [email protected]). Publisher Item Identifier S 0018-9480(99)08445-8.

Fig. 1. OMT operation within the transmit/receive (Tx/Rx) chain.

[2], [3], or for publications in microwave and antenna magazines. There are only a few papers concerning OMT’s, but nearly all of them present sophisticated design for special applications [4], [5]. Several design variants of OMT’s may be found in [1] and the selection of a particular OMT structure mainly depends on the extension of the operating frequency band and on the power-handling limitations. Moreover, whenever a full-wave computer-aided design (CAD) and optimization of the OMT is of interest, it is convenient to select a structure that is amenable to fast computer analysis. In fact, several proposed designs are not particularly suited for analysis since it is necessary to employ general codes (such as HP HFSS) which, unfortunately, are very computer intensive. An interesting remedy to this situation has recently been introduced in [6] and [7]; however, for the OMT case, specific developments are feasible. From the previous discussion, it is apparent that, although the basic structure of an OMT is well known, there is little information (if any) available on the full-wave analysis and optimization of such structures and, in particular, no fullwave optimization of the whole OMT (i.e., also including the final transition between square and circular waveguides) have been presented thus far. It is, therefore, our purpose to describe a procedure for the full-wave design and optimization of state-of-the-art OMT’s. In this paper, we describe a CAD tool that implements the following significant novelties: 1) it makes use of two different full-wave simulators (this fact enhancing accuracy, efficiency, and robustness with respect to numerical convergence problems); 2) it employs a hierarchy of different topologies depending on the required electrical performances of the OMT; 3) it makes use of a dynamical optimization process, which allows updating the number of considered discontinuities according to the selected objective function. This paper is organized as follows. After a description of the selected OMT structure and its modus operandi, there is a brief

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Fig. 3. Schematic view of the branching-type OMT corresponding to the -breadboard of Fig. 2. It is possible to recognize the four different regions: the stepped transition between the in-line port and square waveguide (region 1); the side-port region, i.e., the stepped right-angle bend (region 2); the T-junction in the square waveguide (region 3), which may also include a septum or an image load; and the transition between the square waveguide and circular waveguide (region 4).

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Fig. 2. Examples of manufactured OMT’s operating at side), (top, right-hand side), and band (bottom).

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description of the electromagnetic codes used for the analysis. The numerical convergence of the considered techniques is then investigated by referring to a specifically developed test case. The dynamical optimization procedure suitable for such component is illustrated in Section V. Finally, in Section VI, a comparison of the theoretical and experimental results is presented and issues concerning sensitivity and manufacturability are discussed. II. OMT OPERATING PRINCIPLES The considered OMT essentially consists of a longitudinal stepped transformer, which connects the in-line rectangular waveguide to the square waveguide supporting the two polarizations. This section is combined with a branching waveguide (side port) through a slot centered on the square waveguide wall. The side port is placed in such a way as to allow the propagation of the horizontally polarized field only, while the in-line port permits the propagation of the vertically polarized field only. Finally, since the square waveguide is used to feed a circular horn, a matched transition between a square and

circular waveguide is considered. Actual structures are shown in Fig. 2 and a schematic is illustrated in Fig. 3. From the electrical viewpoint, the OMT operating principles are fairly apparent. The two polarizations present on the feed and circular waveguide may travel to the square waveguide section. For the horizontal polarization, the in-line port is a waveguide below cutoff and, consequently, this field is reflected and coupled to the branching waveguide port only. On the other hand, the vertically polarized field can propagate from the square waveguide to the in-line waveguide, while it is below cutoff at the side port. Note that the lack of symmetry of the structure may excite undesired modes: in fact, the horn specifications in terms of the cross-polar level assume that, at its input port, only the mode is present; unfortunately, within the fundamental is also not under cutoff and, required frequency band the if present, may affect the feed cross-polar performance [8]. Therefore, a particular attention in the OMT design is devoted in order to prevent such excitation; to this end, it is expedient to reduce the waveguide size in the region close to the coupling slot in the branching port. In this way, the unwanted modes remain well below cutoff for a sufficient length of guide and cause no appreciable degradation at the output port. It is worthwhile to note that the classical solution used in ) order to prevent the excitation of higher order modes ( consists of a balanced structure with two symmetric branches derived in the -plane. In such an approach, the higher order mode suppression is obtained at the cost of a significant increase in complexity of the mechanical structure [9]. It is to be noted that a single-branched unbalanced structure can be made quasi-balanced by introducing in the branching region, on the opposite side with respect to the slot, an image load [8]. Nevertheless, this requires the use of an X-junction in place of a T-junction, with consequent increase of computer time and manufacturing cost. As such, this solution needs to be employed only when the simpler topologies fail to deliver the expected performances.

LUDOVICO et al.: CAD AND OPTIMIZATION OF COMPACT OMT’S

From the functional viewpoint, the OMT can be separated into four different regions of space, as specified in the caption of Fig. 3. For future use in the optimization routine, it is important to consider how the relevant design parameters, i.e., the return loss at the in-line and side port and the generation , are related to the different regions. It is fairly of the apparent that the return loss of the in-line port is strictly related to the design of region 1 and, to some extent, also to the design of regions 3 and 4; on the other hand, the structure of region 2 is not influent on this parameter. Similarly, the return loss of the side port is strictly dependent on the design of the bend (region 2) and of the coupling iris with the square waveguide (region 3); the square to circular transition also has some effect, while the stepped transition of region 1 does not affect generation the return loss of the side port. Finally, the in the output circular waveguide is mainly dependent on the electrical symmetry of regions 3 and 4. III. ELECTROMAGNETIC MODELING OF OMT’s Several different numerical techniques are available for the full-wave analysis of OMT’s, such as finite or boundary element or finite differences, both in time and frequency domain, to cite just a few. However, the structure selected in the previous section presents a fairly regular geometry allowing the use of modal techniques that have proven to be both accurate and efficient, the latter qualities being of foremost importance for the computer-aided OMT optimization. The following two different modal techniques have been used for the OMT design, namely: 1) the generalized scattering matrix (GSM) description of each discontinuity; in this case, the necessary “building blocks” are: the double step discontinuity for rectangular waveguides, the right-angle bend, the T-junction, the X-junction, and the rectangular to circular waveguide transition (the GSM description of the latter discontinuities is well known and does not require further discussion) and 2) the so-called “box” approach, based on the generalized admittance matrix (GAM) representation [10]–[12] of a region of space of finite dimension. In particular, in the latter case, a general code for the description of a parallelepipedal box with rectangular apertures on the various faces of the box has been developed. It is noted that, in this description, an aperture may also extend to the entire face of the box or, on the contrary, a face may also be without apertures. Appropriate dyadic Green’s functions have been selected in order to represent the field inside the cavity by rapidly convergent expansions; modal basis functions have been used for describing the field on the apertures. The various admittance representations of different regions are not cascaded together, rather they are assembled according to circuit topology in an overall matrix using the multiport connection method and separating internal and external ports. This methodology has proven effective for the use of the adjoint network method and has been discussed elsewhere [13]–[16]. Details on the relevant parametrization used, and several other features of this approach, are reported in [12]. Note that the “box” approach is extremely versatile; in fact, with the exception of the last discontinuity between a square

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Fig. 4. Three-dimensional view of the coupling slot used for testing the GSM and GAM convergence. This double T-junction has been built and measured using WR75 as input waveguides. The slot is vertically centered and its dimensions are 23.461 mm and 2.193 mm; it is placed at 2 mm from the short circuit.

and a circular waveguide, the entire OMT can be described in terms of several parallelepipedal boxes connected together via apertures. The latter description is provided via a data file that specifies the box dimensions, the number, position, and dimensions of the various apertures and the relevant parameters of interest for the simulation (e.g., the frequency range, etc.). It is also worthwhile to point out that, in this approach, once the code for the single aperture has been developed, it does not necessitate further changes for analyzing different types of discontinuities or components, provided that they can be divided into parallelepipedal boxes connected together through rectangular apertures. The use of the above two methods for the OMT CAD provides interesting information concerning both accuracy and efficiency. With respect to accuracy, we have noted that the two methods generate similar results for a well-chosen parametrization of the problem. Since the two methods employ different techniques, i.e., modal expansions for the GSM and dyadic Green’s functions for the “box” approach, coincidence of results allows an increased confidence of the designer in the theoretical response. However, as for all numerical methods, it is crucial to satisfy a well-chosen problem parametrization; in particular, the number of modes to be selected must be investigated carefully in order to obtain the required accuracy while keeping the numerical burden to a minimum. This last point is particularly important in view of the optimization process necessary for the component design. In the OMT case, the most sensitive discontinuity is the coupling slot relating the side port to the inline section. The accurate modeling of such discontinuity with respect to numerical convergence is investigated in Section IV. IV. NUMERICAL CONVERGENCE OF THE ELECTROMAGNETIC SIMULATORS During a preliminary design, it has been found that, in order to achieve a significant bandwidth (i.e., greater than 30%), it is advantageous to employ a slot that is relatively narrow in the vertical dimension while being quite wide in the horizontal dimension, as illustrated in Fig. 4. Interestingly enough, while the problem of relative convergence has been carefully examined for discontinuities on the longitudinal axis

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Fig. 5. Reflection coefficient of the double T, obtained via the GSM and GAM approaches. We have used 54 TE modes in the input (reference) waveguide.

(see, e.g., references contained in [17]), it has received modest attention for discontinuities placed parallel to the propagation direction, as in the OMT case. Therefore, in order to test the convergence properties of the theoretical solutions, we have built and measured the structure shown in Fig. 4, containing the most critical discontinuities found in the OMT design. The structure is essentially composed of two T-junctions (with a short-circuited branch), which are coupled via a narrow slot. This structure has been simulated with the two numerical methods mentioned above, i.e., GSM and GAM, and the relative results are illustrated below.

(b) Fig. 6. Two different segmentations for the double T-junction and their network equivalent. The GAM code evaluates for each region the corresponding admittance matrix. Note that the segmentation (b) also corresponds to the one used with the GSM approach.

A. Comparison Between the GSM and GAM Results In order to establish the number of modes necessary to represent the field on the discontinuities, we have used the well-known criterion that relates this number to the geometrical dimensions; it is sometimes referred to as a spectral criterion since the same spatial frequency is used throughout the entire structure. In Fig. 5, we have considered the same number of modes in the input reference waveguide and we have analyzed the structure with both the GSM and GAM, obtaining a very close agreement between the two methods. However, no information is provided on the convergence properties, which are now discussed separately for the GAM and GSM. B. GAM Numerical Convergence The GAM formulation essentially presents the same type of convergence properties of the GSM formulation but, in addition, allows dividing the same physical structure into different segmentations, as illustrated in Fig. 6. All different segmentations should generate the same results when the modal set allows an accurate field representation, otherwise we may obtain different results depending on the adopted segmentation of the structure. This behavior is illustrated in Fig. 7, with the experimental results reported for comparison.

Fig. 7. Reflection coefficient of the double T-junction as computed with the GAM. The analysis has been carried out by considering 36 TE modes in the reference waveguide. It is noted that the numerical sensitivity with respect to the different segmentations suggests that the number of modes considered are not yet enough for generating accurate results. Solid line: measurement, dot-dashed line: first segmentation [see Fig. 6(a)], dashed line: second segmentation [see Fig. 6(b)].

Fortunately, with the GAM, it is possible to take advantage of the different segmentations in order to test the convergence of the results; i.e., when a sufficiently high number of modes is selected, different segmentations provide the same result.


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Fig. 8. Building blocks (T-junctions and short circuits) used for the analysis of the double T-junction with the GSM. The small rectangular boxes represent waveguide sections.

Fig. 10. Comparison of the measured and computed reflection coefficient of the double T-junction at 14 GHz as a function of the number of modes used in the GSM analysis. The latter is chosen according to the standard spectral criterion. The unstable convergence behavior suggests possible difficulties in evaluating the number of modes required for the component accurate analysis.

Fig. 9. Reflection coefficient for the double T-junction as obtained from the GSM analyses. The curves obtained for different modal sets are compared with the experimental values. It is possible to observe that convergence seems to be achieved when 55 modes are considered in the reference waveguide. Solid line: measurement, dashed line: 36 TE modes, dot-dashed line: 55 TE modes.

C. GSM Numerical Convergence In the case of the GSM, the structure is divided as illustrated in Fig. 8, and we cannot take advantage of the different segmentations as previously discussed. The choice of the number of modes used in the simulation has an impact on the result accuracy, as illustrated in Fig. 9. In the GSM case, it is of relevance to establish some criteria for testing the numerical convergence. In Figs. 10 and 11, we have compared the computed and measured return loss for different numbers of modes in the reference waveguide. In particular, in Fig. 10, we have considered the spectral criterion that is currently employed in longitudinal discontinuities, while in Fig. 11, we have used a different criterion. This criterion, based on the consideration that the representation of the fields is more critical in the branching region, requires the use of an increased number of modes in the T-junction. As it is apparent from Fig. 11, this last criterion provides a significantly improved convergence. Having investigated the numerical properties relative to the electromagnetic simulators for the structure under consideration, we are now in a position to apply the latter for a reliable optimization of the OMT, as described in Section V. V. OMT DESIGN

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OPTIMIZATION

It is convenient to divide the design of the OMT into the following three main steps:

Fig. 11. Comparison of the measured and computed reflection coefficient of the double T-junction at 14 GHz as a function of the number of modes used in the GSM analysis. A new criterion is used this time for the selection of the modal sets. It is noted that a significant improvement of the convergence behavior has been attained.

1) selection of the OMT structure; 2) choice of the geometrical dimensions of a “minimal” (i.e., with very few discontinuities) OMT, which corresponds to the choice of the starting point for the optimization procedure; 3) dynamical optimization of the entire OMT. The above three items are now described with more details in Sections V-A, V-B, and V-C. A. Selection of the OMT Structure As already mentioned in Section I, the selection of a particular OMT structure depends on its requirements, which typically consider the frequency band of interest, the polarization purity (as discussed in Section II, significant attention is devoted to mode and levels of less than avoiding excitation of the 40 dB are a common requirement), the return loss at the in-line and side ports.

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The preliminary step in order to achieve an efficient CAD system has been the identification of a hierarchy of OMT arrangements flexible enough to be suitable for several different applications. Generally, narrow-band OMT’s have a single branch structure, thus being mechanically compact and easy to manufacture. On the contrary, wide-band or high-power OMT’s require a dual-branch balanced structure and, consequently, they present a remarkable increase in mechanical complexity [1]. Interestingly, computer simulations show that, in several instances, the operating bandwidth can be extended, yet maintaining a single branching structure, by inserting a metallic septum along the main branch. The selected topology of the coupling region is characterized by a vertical plane of symmetry that, theoretically, should provide perfect isolation between the two polarizations. As a matter of fact, isolation directly depends on the mechanical manufacturing accuracy. By considering the latter in a tolerance analysis, it is possible to verify whether the proposed design satisfies specifications or if it requires further adjustments. On the other hand, since a single branch OMT does not possess a second symmetry plane, higher order modes may appear and reduce the operating bandwidth. This drawback can be well-compensated by inserting a mirror , in the form of a waveguide discontinuity load for on the opposite side of the coupling slot [5]. The described inclusion of a metallic septum and/or a mirror load represents an option available during the optimization procedure. Finally, the requirement for a minimum return loss at input ports is met by optimizing the stepped transformers.

Fig. 12. Theoretical and experimental return loss at the in-line input port (top) and at side input port (bottom) of the L-band OMT.

B. Choice of the Initial Dimensions In order to select the starting point for the optimization, we separately analyze (and optimize) the four subregions introduced in Section II (see also Fig. 3). In particular, we start by optimizing the width of the input square waveguide of the T-junction and the relative iris discontinuity on the side port. Typically, the latter is selected as a tradeoff between two contrasting issues: the iris aperture should be fairly large for not blocking the side-port polarization; on the contrary, its dimensions should be as small as possible for not perturbing the in-line polarization. In any case, the width of the square T input and iris size are optimized with respect to the return loss of both the side and in-line ports. After finding these dimensions, it is possible to consider the transition between the square waveguide of T-junction and output circular waveguide, which is generally specified since it is provided from the feed design. Finally, we consider region 1, where we introduce some step discontinuities in order to optimize the junction between the input in-line rectangular waveguide and output square waveguide. A similar optimization is also carried out in region 2, but this time the bend discontinuity is also considered. At this point, a set of geometrical dimensions has been generated and it is possible to start the optimization of the entire OMT. C. Dynamical Optimization of the OMT The objective function to be optimized takes into account three different contributions that arise, respectively, from the

return loss of both the in-line and side input ports and the mode, which, if excited, can necessity to control the propagate in the circular waveguide. Noticeably, the various contributions are strictly related to the different regions of the OMT. As discussed in Section II, the return loss of the in-line waveguide depends severely on the number of steps in region 1, and only to a minor extent does it depend on the discontinuities present in regions 3 and 4. In addition, there is only a very weak relationship between the geometry of region 2 and the return loss in the in-line port. Similar considerations also hold for the return loss of the side port and polarization purity. According to the above functional dependence, it is also possible to devise a strategy for the efficient dynamical optimization of the OMT. In fact, it is expedient to separately examine the contributions arising from the return loss of the in-line port, the return loss of mode generated. By the side port, and the amount of considering the larger of these contributions, we find what region needs further improvement; typically a further step discontinuity is added to this region, increasing the number of degrees of freedom for the optimization. The code then performs a full-wave optimization on that particular region and, subsequently, the entire OMT structure is optimized. If specifications are met, then there is no need for introducing further discontinuities; otherwise, the process is repeated and further discontinuities are introduced in the appropriate region.


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gated carefully with the help of a measured test case, leading to a new convergence criterion for discontinuities parallel to the propagation direction. In addition, a procedure for the dynamical computer-aided optimization of OMT has been illustrated and used for designing several OMT’s operating in , , and frequency bands. The component design, entirely carried out at computer level, has demonstrated significant advantages in terms of development times and no need of post-manufacturing adjustments. The very satisfactory agreement between experimental and theoretical results further confirms the validity of the proposed technique. ACKNOWLEDGMENT Fig. 13. Theoretical return loss at the in-line input port and at side input -band OMT (30% bandwidth, 10.75–14.5 GHz). port of a

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In this way, by starting from the “minimal” design, the structure is dynamically updated and further discontinuities (waveguide steps) are introduced until the specifications are met. VI. EXPERIMENTAL AND THEORETICAL RESULTS By using the above optimization procedure, several OMT’s operating over different bandwidths have been designed and built: in particular, in [18], an OMT operating at the 17–18.5-GHz frequency band has been described, while in the following, we discuss OMT’s operating at -, -, and -bands. The two OMT breadboards, operating in the - and bands, respectively, are shown in Fig. 2; these components have been fabricated in aluminum, using numerically controlled milling and spark erosion machines to achieve an accuracy within 0.04 mm. Return-loss computation and measurement for both in-line and side inputs of -band OMT are presented in Fig. 12. The measured return loss is better than 30 dB over the whole operating frequency band (1.5–1.75 GHz) and exhibits a good agreement with computer simulations, thus confirming the efficiency and accuracy of the proposed electromagnetic model. The measured isolation between vertical and horizontal polarization is higher than 55 dB. The electrical performance of -band OMT, not reported here for brevity, is similar. The manufactured components, which have been designed with the metallic septum option, make available a fractional bandwidth of approximately 15%. As a final example we have considered the case of an OMT operating over a 30% bandwidth (10.75–14.5 GHz). Fig. 13 shows the return-loss computation for this component, which includes both metallic septum and mirror load options [8]. VII. CONCLUSIONS A hybrid CAD technique, which employs the standard GSM description, and a new one based on the three-dimensional admittance representation has been successfully used for the efficient and reliable analysis of compact OMT’s. Numerical convergence of the considered techniques has been investi-

The authors wish to thank Alcatel Space, Cannes, France, and G. Manara and A. Monorchio, Dipartimento di Ingegneria dell’Informazione, Universit`a di Pisa, Pisa, Italy, for their continuing encouragement and many fruitful discussions. REFERENCES [1] J. Uher, J. Bornemann, and U. Rosenberg, Waveguide Components for Antenna Feed Systems. Norwood, MA: Artech House, 1993. [2] G. Meinke, Taschenbuch der Hochfrequenztechnik. Berlin, Germany: Springer-Verlag, 1986. [3] A. F. Harvey, Microwave Engineering. New York: Academic, 1963. [4] H. Schlegel and W. D. Fowler, “The ortho-mode transducer offers a key to polarization diversity in EW systems,” Microwave Syst. News, pp. 65–70, Sept. 1984. [5] S. J. Skinner and G. L. James, “Wide-band ortho-mode transducers,” IEEE Trans. Microwave Theory Tech., vol. 39, pp. 294–300, Feb. 1991. [6] J. Bandler, R. Biernacki, S. Chen, and D. Omeragic, “Space mapping optimization of waveguide filters using finite element and mode-matching electromagnetic simulators,” in IEEE MTT-S Int. Microwave Symp. Dig., June 1997, pp. 635–638. [7] J. Bandler, R. Biernacki, and S. Chen, “Fully automated space mapping optimization of 3d structures,” in IEEE MTT-S Int. Microwave Symp. Dig., June 1996, pp. 753–756. [8] W. Steffe, “A novel compact OMJ for -band Intelsat applications,” in IEEE AP-S Int. Symp. Dig., vol. 1, June 1995, pp. 152–155. [9] G. F. Cazzatello, M. Barbiero, G. Figlia, and C. G. M. V. Klooster, “A 38% bandwidth circularly polarized feed for radio astronomy applications,” in Proc. JINA, Nice, France, Nov. 1996, pp. 494–506. [10] F. Alessandri, M. Mongiardo, and R. Sorrentino, “Computer-aided design of beam forming networks for modern satellite antennas,” IEEE Trans. Microwave Theory Tech., vol. 40, pp. 117–1127, June 1992. , “A technique for the full-wave automatic synthesis of wave[11] guide components: Application to fixed phase shifters,” IEEE Trans. Microwave Theory Tech., vol. 40, pp. 1484–1495, July 1992. [12] G. Zarba and M. Mongiardo, “Un metodo generale di analisi elettromagnetica 3D per strutture in guida d’onda metallica composte da risonatori di sezione rettangolare,” CSELT, Turin, Italy, Int. Rep., pp. 1–56, Dec. 1997. [13] F. Alessandri, M. Mongiardo, and R. Sorrentino, “New efficient full wave optimization of microwave circuits by the adjoint network method,” IEEE Microwave Guided Wave Lett., vol. 3, pp. 414–416, Nov. 1993. [14] F. Alessandri, M. Dionigi, M. Mongiardo, and R. Sorrentino, “Efficient full-wave automated design and yield analysis of waveguide components,” Int. J. RF Microwave Computer-Aided Eng. (Special Issue), vol. 8, no. 3, pp. 200–207, May 1998. [15] M. Mongiardo and R. Ravanelli, “Automated design of corrugated feeds by the adjoint network method,” IEEE Trans Microwave Theory Tech., vol. 45, pp. 787–793, May 1997. [16] M. Mongiardo, “Sensitivity evaluations of microwave filters by the adjoint network method,” presented at the IEEE MTT-S Int. Microwave Workshop, Denver, CO, June 1997. [17] R. Sorrentino, M. Mongiardo, F. Alessandri, and G. Schiavon, “An investigation on the numerical properties of the mode-matching technique,” Int. J. Numerical Modeling, vol. 4, pp. 19–43, 1991.

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[18] B. Piovano, G. Zarba, G. Bertin, L. Accatino, and M. Mongiardo, “CAD and optimization of compact ortho-mode transducers,” in IEEE MTT-S Int. Microwave Symp. Dig., June 1999, pp. 257–260.

Michele Ludovico was born in Montevarchi, Italy, in 1971. He received the Doctor degree in electronic engineering from the University of Pisa, Pisa, Italy, in 1997. In 1999, he joined the Microwave Department, Centro Studi e Laboratori Telecomunicazioni S.p.A. (CSELT), Turin, Italy, where his current activity is focused on the design and implementation of highperformance waveguide components for antenna feed systems, in particular filters, diplexers, and OMT’s.

Bruno Piovano (A’92) was born in Turin, Italy, in 1953. He received the Doctor degree in electronic engineering from the Polytechnic School of Turin, Turin, Italy, in 1978. In 1980, he joined the Microwave Department, Centro Studi e Laboratori Telecomunicazioni S.p.A. (CSELT), Turin Italy. He first developed a basic and comprehensive software package for analyzing complex beam-forming networks accounting for the electromagnetic model of subcomponents. He then became a Senior Researcher responsible for the design of waveguide networks in the INTEL-717 “flexible/reconfigurable antenna” study and in the European Space Agency “butler matrix amplifier” and “contoured beam reconfigurable Antenna” studies. His current activities regard the design of high-performance waveguide subsystems, with particular attention to antenna feed networks.

Giorgio Bertin (M’92) was born in Aosta, Italy, in 1956. He received the Doctor degree in electronic engineering from the Polytechnic School of Turin, Turin, Italy, in 1982. In 1983, he joined the Microwave Department, Centro Studi e Laboratori Telecomunicazioni S.p.A. (CSELT), Turin, Italy, where he was first engaged in dielectric oscillator and dielectric-loaded cavity design. His activities then focused on the modeling of microwave discontinuities and the computeraided design of guiding structures, with particular attention devoted to discontinuities between nonstandard waveguides, such as those involving a dielectric loading or the presence of ridges. He currently supervises all activities related to electromagnetic modeling in the Microwave Department.

Giovanna Zarba (A’97–M’97) was born in Turin, Italy, in 1967. She received the Doctor degree in electronic engineering from the Polytechnic School of Turin, Turin, Italy, in 1992. In 1992, he joined the Microwave Department, Centro Studi e Laboratori Telecomunicazioni S.p.A. (CSELT), Turin, Italy, where she was engaged in the design and implementation of microstrip components and networks used in front-end systems. Since 1994, she has been involved in the study of novel waveguide components for beam-forming networks, channel filters, and diplexers and multiplexers for use on board satellites, with attention focused on the development of advanced and efficient electromagnetic models for a design entirely at the computer level of these components.

Luciano Accatino (M’84) was born in Turin, Italy, in 1950. He received the Doctor degree in electronic engineering from the Polytechnic School of Turin, Turin, Italy, in 1973. In 1975, he joined the Centro Studi e Laboratori Telecomunicazioni S.p.A. (CSELT), Turin, Italy, where he was initially engaged in the design of microstrip circuits and components. In 1980, he became involved in the design and development of microwave cavity filters and, subsequently, of various components for beam-forming networks. He then supervised the activities related to filters and waveguide components at CSELT, stimulating a wide application of electromagnetic models to the design of all passive components. Since 1994, he has been Head of the Microwave Department.

Mauro Mongiardo (M’91), for photograph and biography, see this issue, p. 2478.