Design and Control of a Bidirectional Resonant DC& ... - IEEE Xplore

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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 7, JULY 2014

Design and Control of a Bidirectional Resonant DC–DC Converter for Automotive Engine/Battery Hybrid Power Generators Junsung Park, Student Member, IEEE, and Sewan Choi, Senior Member, IEEE

Abstract—In this paper, a bidirectional dc–dc converter is proposed for the automotive engine/battery hybrid power generators. The two-stage bidirectional converter employing a fixed-frequency series loaded resonant converter is designed to be capable of operating under zero-current-switching turn-on and turn-off regardless of voltage and load variation, and hence its magnetic components and EMI filters can be optimized. Also, a new autonomous and seamless bidirectional voltage control method that combines two individual controllers for low-voltage side control and high-voltage side control by introducing a variable current limiter is proposed to provide uninterrupted power to critical ac loads and reduce the size of the dc bus capacitor and the transition time. Experimental results from a 5-kW prototype are provided to validate the proposed concept. Index Terms—Bidirectional dc–dc converter (BDC), hybrid power generator, seamless transition, series loaded resonant converter (SRC), zero-current switching (ZCS).

COSSp COSSs Cr Dd Ts DTs fr fr (eff ) fs1 fs2 IL ILB ILB,H ILB+ ILB+,pk ILB−,pk

NOMENCLATURE Primary side MOSFETs’ output capacitance. Secondary side MOSFETs’ output capacitance. Resonant capacitance (Cr = Cr 1 + Cr 2 ). Dead time of the SRC. On-time duty cycle of the SRC. Resonant frequency. Effective resonant frequency. Switching frequency of the SRC. Switching frequency of the nonisolated converter. Low side current of the SRC. Inductor current of the nonisolated converter. Input to the variable current limiter. Positive limit of the variable current limiter. Peak value of the positive limit of the variable current limiter. Peak value of the negative limit of the variable current limiter.

Manuscript received February 18, 2013; revised May 7, 2013 and June 26, 2013; accepted September 2, 2013. Date of current version February 18, 2014. Recommended for publication by Associate Editor D. Xu. J. Park is with the Power Electronics & Fuel Cell Power Conditioning Laboratory, Department of New Energy Engineering, Seoul National University of Science and Technology (Seoul Tech), Seoul 139-743, Korea (e-mail: [email protected]). S. Choi is with the Department of Electrical and Information Engineering, Seoul National University of Science and Technology (Seoul Tech), Seoul 139743, Korea (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TPEL.2013.2281826

IL m IL r ISH ISL Lk p Lk s Lm Lr M PB PBDC PG Pi PLoad,ac PLoad,dc ttran VH Vi VL VSH VSL

Magnetizing inductor current of the SRC. Resonant current of the SRC. High side switch current of the SRC. Low side switch current of the SRC. Primary side leakage inductance of the SRC. Secondary side leakage inductance of the SRC. Magnetizing inductance of the SRC. Resonant inductance of the SRC. Gain of the SRC (M = VL /Vi ). Input power of the battery. High side input power of the BDC. Output power of the engine generator. High side input power of the SRC. Demanded ac load power. Demanded dc load power. Mode transition time. High side voltage of the BDC. High side voltage of the SRC. Low side voltage of the SRC (battery voltage). High side switch voltage of the SRC. Low side switch voltage of the SRC.

I. INTRODUCTION TANDBY or emergency generators are often used as backup power supplies for buildings, industrial facilities, and power plants in the event of a loss of utility power [1]. In addition, remote power generation for military, industrial, and personal use requires a reliable, compact, and a lightweight power generation system. The diesel generation system has been used as backup power supplies or remote power generators [2]. Since the engine generator may not be able to respond to sudden load changes, energy storage devices should be used along with the engine generator to level out the erratic changes in power balance between the generation and load consumption [3], [4]. Energy storage device is used along with a bidirectional dc– dc converter (BDC) in order to match the voltage level and/or achieve efficient charging and discharging operation [2]. Fig. 1 shows an automotive engine/battery hybrid power generation system. The BDC is located between the high-voltage dc bus and the low-voltage battery which is also connected to the dc loads such as antilock brakes, electric power steering, heated seats, electronic ignition, and HVAC in the vehicle. The dc–ac inverter converts the dc power to ac power to supply the critical ac load in the vehicle such as broadcasting equipment of outside broadcast van and communications equipment of

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PARK AND CHOI: DESIGN AND CONTROL OF A BIDIRECTIONAL RESONANT D–DC CONVERTER

Fig. 2.

Fig. 1.

Automotive engine/battery hybrid power generation system.

tactical vehicle. The ac–dc converter converts the ac power from the engine generator to the dc power, regulating the high-voltage dc bus [5]. If the engine generator is capable of supplying the total demanded power of ac and dc loads, the ac–dc converter will be able to regulate the high-voltage dc bus, and the BDC will deliver the power from the engine generator to the low-voltage side. If the engine generator is shut down or the total demanded power of the ac and dc loads is greater than the maximum power of the engine generator, high side bus voltage will drop off to a voltage depending on the capacitances of the dc bus capacitor. Then, the BDC is required to take over the regulation duty of the high-voltage dc bus by changing over from VL control (battery charging) to VH control (battery discharging) so that it should be able to deliver power from the battery to the ac load. Therefore, in order to provide uninterrupted power to the critical ac loads and reduce the size and cost of the dc bus capacitor, the transition from VL control to VH control of the BDC should be seamless and as short as possible. This is a crucial performance of the BDC, especially, in the automotive application where electrolytic capacitors cannot be used due to limited lifespan and bulky nature [6]–[9]. A seamless changeover of the control target from VL control to VH control or vice versa has not been discussed so far. The BDC should provide a galvanic isolation and a high step up/down voltage conversion ratio in the application where the low-voltage battery is used. Typical topology candidates with these requirements include half-bridge, full-bridge, and push-pull pulse width modulation (PWM) converters [10], [11], dual active bridge (DAB) converters [12], [13], and two-stage converters [14], [15]. The PWM converters usually necessitate passive or active clamping on the low-voltage side to clamp the surge voltage generated by the leakage inductance of the transformer. The active clamping technique makes the converter not only clamp the surge voltage, but achieve zero-voltageswitching (ZVS) turn-on of all the switches. A drawback of the active clamped PWM converter is high turn-off switching losses [16]. The DAB has a modular and symmetric structure and can achieve ZVS turn-on without auxiliary components. However, the DAB has limited ZVS range and high-circulating currents for applications requiring wide voltage variation. The ripple current of the DAB converters is high and especially problematic in the low-voltage applications [17]. Two-stage converters consist of a nonisolated stage and an isolated stage. Since the nonisolated stage is operated to regulate the voltage and power

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Proposed two-stage BDC.

flow, the isolated stage can be designed with minimum components’ rating. Even though the power is passing through two conversion stages, the two-stage converter could achieve a higher efficiency especially in a wide voltage range application [14], [15]. In this paper, a two-stage BDC is proposed for automotive engine/battery hybrid power generators. The proposed two-stage BDC consists of a nonisolated converter and a fixed-frequency SRC. The SRC is designed to be capable of operating under ZCS turn-on and turn-off regardless of voltage and load variation in both forward and reverse operation. A method of adjusting dead time of the SRC will be presented to minimize the turn-on switching losses associated with energy stored in MOSFET’s output capacitances during the ZCS turn-on process. Also, a new autonomous and seamless bidirectional voltage control strategy is proposed to provide uninterrupted power to the critical ac loads and reduce the size of the dc bus capacitor and the transition time. II. PROPOSED BDC The proposed BDC consists of two power conversion stages: a nonisolated converter and a fixed-frequency SRC, as shown in Fig. 2. Since the SRC is operated at fixed frequency and fixed duty, all components can be designed with minimum voltage and current rating. The nonisolated converter is operated to regulate either high side voltage VH or low side voltage VL according to the demanded load power and availability of the engine generator. Figs. 3 and 4 show key waveforms and operation states of the proposed SRC, respectively. The angular resonant frequency of the resonant circuit can be expressed as 1 ωr = 2πfr = √ Lr · Cr

(1)

where resonant inductance and resonant capacitance can be determined, respectively, by Lr = Lk p +

Lm · n2 Lk s Lm + n2 Lk s

Cr = Cr 1 + Cr 2 .

(2) (3)

It is seen from Figs. 3 and 4 that the low side current iL (=iS L 2 ) at Mode I(t0 −t1 ) becomes purely sinusoidal if the on-time duty cycle is selected such that DTs = 0.5/fr . Then it can be expressed as iL (t) =

πIL ,dc sin ωr t. 2

(4)

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Also, voltage across Lm can be expressed as diL vL m (t) = −n VL + Lk s . dt

(5)

Therefore, from (4) and (5), the magnetizing current at Mode I (t0 −t1 ) can be expressed using iL m (t0 ) = – iL m (t1 ) by nπLk s IL ,dc nπVL nVL − t+ iL m (t) = sin ωr t. (6) 2ωr Lm Lm 2Lm The resonant current can then be obtained using (4) and (6) by nπLk s IL ,dc nπVL nVL πIL ,dc − t+ − iL r (t) = sin ωr t. 2ωr Lm Lm 2Lm 2n (7) Neglecting voltage oscillation after turning ON of SL 2 , the voltage across low side switch SL 1 at Mode I (t0 −t1 ) is expressed as vSL1 (t) = 2VL + Lk s

diL . dt

(8)

The turn-off voltage of low side switch can be obtained by VSL,off = 2VL −

πωr Lk s IL ,dc . 2

(9)

It should be noted that VSL,off should be greater than zero for the proposed operation. Therefore, from (4) and (9) the secondary side leakage inductance should be limited such as Lk s
fr , the SRC behaves like region II operation of LLC converter and is turned OFF with the magnetizing current at turn-off instant. Note that the proposed SRC can be said to be turned OFF with near ZCS since the magnetizing current of the proposed converter is very small compared to LLC converter due to the larger magnetizing inductance. When fr (eff ) < fr , the SRC is operated under inductive region and turned OFF with hard switching. III. PROPOSED CONTROL STRATEGY The high side dc bus is regulated to either 400 V by the ac– dc converter or 380 V by the BDC, respectively, according to the condition of VH . The conventional control of the BDC is in general realized with the two individual controllers of VL control for battery charging and VH control for battery discharging, and therefore may not be able to avoid large transient during the transition from VL control to VH control of the BDC.

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Fig. 7.


Control block diagram of the proposed battery charger.

In this paper, a new autonomous and seamless bidirectional voltage control strategy, as shown in Fig. 7, is proposed to provide uninterrupted power to the critical ac loads and reduce the size of the dc bus capacitor. The two outer loop voltage controllers for VL control and VH control are combined by vari∗ is automatically able current limiter (VCL) whose output ILB selected to be either ILB,H , the output of the high side voltage controller, or ILB+ , the positive limit of VCL which varies with the output of the low side voltage controller. This makes it possible to share inner-loop current controller, resulting in autonomous and seamless transition from VL control (charging mode) to VH control (discharging mode), and vice versa. The peak values of the positive and negative limits, ILB−,pk and ILB+,pk , of the VCL are determined by ILB+,pk = ILB−,pk =

Pi . Vi

(17)

According to C-rate of the battery used, ILB+,pk may be chosen smaller than (17). ILB+ varies with magnitude of VL , while ILB− is always fixed at ILB−,pk . The antiwindup is used to prevent the saturation of the controllers. For the sake of simplicity, it is assumed that the dc-load is constant and all the power losses of the ac–dc converter, the dc–ac inverter, and the BDC in Fig. 1 are neglected. A. Transition From VL Control to VH Control Figs. 8 and 9 show PSIM simulation waveforms and operation states of the VCL, respectively, for illustration of the operating principle of the proposed bidirectional control strategy for transition from VL control to VH control. Mode I: Assume that the battery has already been fully charged. The engine generator is supplying the ac and dc loads during this mode. VH is regulated to 400 V by the ac–dc converter, and the reference voltage VH∗ of the BDC is set at 380 V. Since the high side voltage controller GH (s) is saturated, the ∗ of the BDC is determined by IL B + which reference current ILB is the same as ILoad,dc /M, the dc load current, as shown in Fig. 9(a). Mode II: This begins when the ac load increases and the sum of the ac and dc loads is greater than PG ,m ax , the maximum power that can be produced by the generator. Then, the ac–dc converter is not able to regulate the dc bus, and VH drops off

Fig. 8. Simulation waveform of the proposed bidirectional voltage control for seamless transition from V L control to V H control. TABLE I SUMMARY OF THE PROPOSED BIDIRECTIONAL VOLTAGE CONTROL FOR SEAMLESS TRANSITION FROM V L CONTROL TO V H CONTROL


Fig. 9.

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Operation states of the variable current limiter for seamless transition from V L control to V H control: (a) Mode I, (b) Mode II, and (c) Mode III.

TABLE II SUMMARY OF THE PROPOSED BIDIRECTIONAL VOLTAGE CONTROL FOR SEAMLESS TRANSITION FROM V H CONTROL TO V L CONTROL

B. Transition From VH Control to VL Control

Fig. 10. Simulation waveform of the proposed bidirectional voltage control for seamless transition from V H control to V L control. ∗ from 400 V, releasing the saturation of GH (s), which makes ILB ∗ be changed to ILB,H , as shown in Fig. 8. ILB decreases, changes its sign, and continuously increases, as shown in Fig. 9(b). This means that the BDC starts to discharge the battery and regulate VH to 380 V. As the battery voltage decreases, ILB+ which is the output of the low side voltage controller GL (s) increases up to ILB+,pk . This is the end of the mode. ∗ is fixed at a constant value since the ac load Mode III: ILB is constant, as shown in Fig. 9(c). The BDC keeps discharging the battery and regulating VH to 380 V. ILB+ is kept at ILB+,pk . The characteristics of the proposed bidirectional voltage control for seamless transition from VL control to VH control is summarized in Table I.

Figs. 10 and 11 show PSIM simulation waveforms and operation states of the VCL, respectively, for illustration of the operating principle of the proposed bidirectional control strategy for transition from VH control to VL control. Mode I: This mode is identical to Mode III of Section III-A. The BDC is discharging the battery and regulating VH to 380 V. ∗ ILB is determined by ILB,H , and ILB+ is the same as ILB+,pk . Mode II: This mode begins when the ac load decreases and the sum of the ac and dc loads becomes smaller than PG ,m ax . This makes the ac–dc converter capable of regulating VH , recov∗ (= ILB,H ) decreases and ering it back to 400 V. Therefore, ILB changes its sign, meaning that the BDC is able to regulate VL to 28 V, and continuously increases until it reaches ILB+,pk , ∗ is determined by ILB+ as shown in Fig. 11(b). Now, ILB (=ILB+,pk ) since the high side voltage controller GH (s) is saturated. Then, the BDC starts to charge the battery with a constant current of IB ,CC which is determined by IB ,CC /M = ILB+,pk − ILoad,dc /M.

(18)

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Fig. 11.


Operation states of the variable current limiter for seamless transition from V H control to V L control: (a) Mode I, (b) Mode II, and (c) Mode III.

Fig. 13. Undershoot voltage and transition time according to total high side capacitance C H , to t in the simulation of Fig. 12. TABLE III PARAMETERS OF THE PROPOSED BDC

Fig. 12. Simulation waveform of mode transition: (a) conventional control and (b) proposed control.

Mode III: When the battery voltage VL gets close to VL∗ , the ∗ which is determined by ILB+ starts to reference current ILB decrease, as shown in Fig. 11(c). During this mode, the BDC charges the battery with a constant voltage of VL∗ . The characteristics of the proposed bidirectional voltage control for seamless transition from VH control to VL control are summarized in Table II.


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Fig. 14. Experimental waveforms of the charging mode: (a) inductor current IL B , switch voltages V S B , 1 and V S B , 2 of the nonisolated converter, (b) primary current Ip ri , high side switch voltages V S H , 1 and V S H , 2 of the SRC, and (c) primary current Ip ri , low side switch voltages V S L , 1 and V S L , 2 of the SRC.

Fig. 15. Experimental waveforms of the discharging mode: (a) inductor current IL B , switch voltages V S B , 1 and V S B , 2 of the nonisolated converter, (b) primary current Ip ri , high side switch voltages V S H , 1 and V S H , 2 of the SRC, and (c) primary current Ip ri , low side switch voltages V S L , 1 and V S L , 2 of the SRC.

Fig. 16. Experimental waveforms of transition from V L control for charging to V H control for discharging: (a) high side voltage V H , low side voltage VL , and inductor current IL B , (b) extended waveforms of inductor current IL B .

Fig. 17. Experimental waveforms of transition from V H control for discharging to V L control for charging (a) high side voltage V H , low side voltage V L , and inductor current IL B , (b) extended waveforms of inductor current IL B .

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as shown in Fig. 12(b). Fig. 13 shows the undershoot voltage and transition time according to CH ,tot in this simulation. The undershoot voltage of the proposed control method is smaller than its limit value without regards to CH ,tot . It is seen from Fig. 13 that the transition time of the proposed control method is 1.5 to 2.5 times faster than that of the conventional control method in the region of CH ,tot > 1000 μF where the undershoot voltage of the conventional method is smaller than the limit value.

Fig. 18. Measured efficiencies of the proposed BDC including gate drive and control circuit losses.

Fig. 19.

Photograph of the proposed BDC prototype.

C. Performance Comparison of the Proposed and Conventional Control Methods Fig. 12 shows the simulation waveforms of the conventional control methods [22], [23] and the proposed control method. VH was initially being regulated to 400 V by the ac–dc converter and is changed to be regulated to 380 V by the BDC when the engine generator is shut down. Considering proper operation of the dc– ac inverter, the absolute minimum input voltage Vabsolute,m in of the dc–ac inverter is restricted to higher than 365 V. In the conventional control, the direction and amount of the energy flow between the battery and the high side voltage bus is determined by comparing the bus voltage level VH with threshold values and changing over from a high side voltage controller to a low side voltage controller, or vice versa. As shown in Fig. 12(a), the BDC starts regulating VH when decreased, VH becomes smaller than the threshold voltage Vth,dischar (= 370 V) for mode transition from charging to discharging, and therefore, the transition time ttran is quite long. In order to reduce the transition time, total high side capacitor CH ,tot can be reduced, but this may increase the undershoot voltage, making the system unstable, as shown in Fig. 12(a). Therefore, in this example CH ,tot should be selected to be larger than 1 mF. Furthermore, sudden changeover of the controller could cause a large inductor current overshoot, as shown in Fig. 12(a), which may result in damage to components of the BDC. In the proposed control method, owing to the VCL, the BDC starts regulating VH from the moment when VH drops off from 400 V. This leads to significantly reduced undershoot voltage of VH and overshoot of inductor current, resulting in reduced ttran ,

IV. EXPERIMENTAL RESULTS Parameters of the proposed BDC are listed in Table III. Figs. 14 and 15 show the key experimental waveforms of the charging and discharging modes at full load, respectively. As we can see from Fig. 14(b), (c) and Fig. 15(b), (c), all switches of the SRC are being turned ON and OFF with ZCS in both charging and discharging modes. In fact, all switches of the SRC are always turned ON and OFF with the ZCS without regard to voltage and load variations. Figs. 16 and 17 show the experimental waveforms of the mode transition. A 24 V/100 Ah lead acid battery was used at the lowvoltage side. Fig. 16(a) shows that the BDC is regulating VL to charge the battery, and VH is regulated to 400 V by the ac–dc converter. When the engine generator is shut down, VH drops but is recovered to 380 V since the BDC changes over to VH control to discharge the battery. It is seen from Fig. 16(b) that there are no transient current surges during the transition from VL control to VH control. Fig. 17(a) shows that VH is being regulated to 380 V by the BDC. When the engine generator restarts, VH is recovered back to 400 V by the ac–dc converter, which makes the BDC to start regulating VL , charging the battery. It is also seen from Fig. 17(b) that there are also no transient current surges during the transition from VH control to VL control. The efficiency of the proposed BDC including gate drive and control circuit losses is measured by YOKOGAWA WT3000 and is shown in Fig. 18. The maximum efficiencies are 95.13% at 1.3 kW in charging mode and 95.08% at 1.5 kW in discharging mode, respectively. Fig. 19 shows the photograph of the proposed BDC prototype. V. CONCLUSION This paper proposes a BDC for automotive engine/battery hybrid power generators. The features of the proposed BDC are as follows: 1) The proposed topology preserves the advantages of the two-stage dc–dc converter: a) The switching method is simple in that voltage regulation and mode transition are carried out only by the nonisolated converter. b) All components’ ratings of the isolated converter are optimized. 2) Small Lr can be used since the proposed SRC is not used for regulation, which leads to the following advantages: a) The SRC has very small gain variation according to load variation, and therefore the proposed BDC can be designed for wider voltage range. b) The SRC is less sensitive to the resonant component tolerances, and therefore suitable


for high volume manufacturing. c) Small Lr can be easily embedded in the transformer. 3) The proposed SRC is capable of achieving ZCS turn-on and turn-off regardless of voltage and load variation. A method of adjusting dead time of the SRC has been presented to minimize the switch turn-on losses associated with energy stored in MOSFET’s output capacitances during the ZCS turn-on process. 4) An autonomous and seamless bidirectional voltage control method with a variable current limiter has been proposed to provide uninterrupted power to critical ac loads and reduce the size of the dc bus capacitor. Experimental results from a 5-kW prototype were provided to validate the proposed concept. The maximum efficiencies including gate drive and control circuit losses are 95.13% at 1.3 kW in charging mode and 95.08% at 1.5 kW in discharging mode, respectively. REFERENCES [1] P. Famouri, W. R. Cawthorne, N. Clark, S. Nandkumar, C. Atkinson, R. Atkinson, T. McDaniel, and S. Petreanu, “Design and testing of a novel linear alternator and engine system for remote electrical power generation,” in Proc. IEEE Power Eng. Soc. Winter Meeting, Jan. 31–Feb. 4, 1999, pp. 108–112. [2] Z. Chen and Y. Hu, “A hybrid generation system using variable speed wind turbines and diesel units,” in Proc. IEEE 29th Annu. Conf. Ind. Electron. Soc., Nov. 2–6, 2003, pp. 2729–2734. [3] E. Muljadi and T. J. Bialasiewicz, “Hybrid power system with a controlled energy storage,” in Proc. IEEE 29th Annu. Conf. Ind. Electron. Soc., Nov. 2–6, 2003, pp. 1296–1301. [4] L. Wang and D. Lee, “Load-tracking performance of an autonomous SOFC-based hybrid power generation energy storage system,” IEEE Trans. Energy Convers., vol. 25, no. 1, pp. 128–139, Mar. 2010. [5] D. Kim and S. Choi, “Load balancing with mobile base stations in tactical information communication networks,” in Proc. IEEE Wireless Commun. Netw. Conf., Mar. 28–31, 2011, pp. 28–31. [6] H. Wen, X. Wen, J. Liu, X. Guo, and F. Zhao, “A low-inductance highfrequency film capacitor for electric vehicles,” in Proc. Int. Conf. Electr. Mach. Syst., Oct. 8–11, 2007, pp. 2046–2050. [7] Y. X. Qin, H. S. H. Chung, D. Y. Lin, and S. Y. R. Hui, “Current source ballast for high power lighting emitting diodes without electrolytic capacitor,” in Proc. IEEE 34th Annu. Conf. Ind. Electron., Nov. 10–13, 2008, pp. 1968–1973. [8] J. Kim and S. Sul, “Resonant link bidirectional power converter. Part II—Application to bidirectional AC motor drive without electrolytic capacitor,” IEEE Trans. Ind. Appl., vol. 10, no. 4, pp. 485–493, Jul. 1995. [9] H. Chae, H. Moon, and J. Lee, “On-board battery charger for PHEV without high-voltage electrolytic capacitor,” Electron. Lett., vol. 46, pp. 1691– 1692, Dec. 2010. [10] L. Rongyuan, A. Pottharst, N. Frohleke, and J. Bocker, “Analysis and design of improved isolated full-bridge bidirectional DC–DC converter,” in Proc. IEEE 35th Annu. Power Electron. Spec. Conf., Jun. 20–25, 2004, pp. 521–526. [11] G. Ma, W. Qu, G. Yu, Y. Liu, N. Liang, and W. Li, “A zero-voltageswitching bidirectional DC–DC converter with state analysis and softswitching-oriented design consideration,” IEEE Trans. Ind. Electron., vol. 56, no. 6, pp. 2174–2184, Jun. 2009. [12] R. W. De Doncker, D. M. Divan, and M. H. Kheraluwala, “A three-phase soft-switched high-power density DC/DC converter for high-power applications,” IEEE Trans. Ind. Appl., vol. 27, no. 1, pp. 63–73, Jan. /Feb. 1991. [13] F. Krismer and J. W. Kolar, “Efficiency-optimized high-current dual active bridge converter for automotive applications,” IEEE Trans. Ind. Electron., vol. 59, no. 7, pp. 2745–2760, Jul. 2012.

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Junsung Park (S’13) received the B.S. and M.S. degrees from the Seoul National University of Science and Technology (Seoul Tech), Seoul, Korea, in 2009 and 2011, respectively. He is currently working toward the Ph.D. degree at the Power Electronics & Fuel Cell Power Conditioning Laboratory, Seoul Tech. His research interests include high-power dc–dc converter for electric vehicles and renewable energy systems.

Sewan Choi (S’92–M’96–SM’04) received the B.S. degree in electronic engineering from Inha University, Incheon, Korea, in 1985, and the M.S. and Ph.D. degrees in electrical engineering from Texas A&M University, College Station, TX, USA, in 1992 and 1995, respectively. From 1985 to 1990, he was with Daewoo Heavy Industries as a Research Engineer. From 1996 to 1997, he was a Principal Research Engineer at Samsung Electro-Mechanics Co., Korea. In 1997, he joined the Department of Electrical and Information Engineering, Seoul National University of Science and Technology (Seoul Tech), Seoul, Korea, where he is currently a Professor. His research interests include power conversion technologies for renewable energy systems and dc–dc converters and battery chargers for electric vehicles. Dr. Choi is an Associate Editor of the IEEE TRANSACTIONS ON POWER ELECTRONICS and IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS.