Electronic interface for differential pressure sensor - IEEE Xplore

Electronic interface for differential pressure sensor Michal Pavlik, Radimir Vrba, Pavel Steffan Dept. of Microelectronics BUT FEKT, Údolní 53, 602 00 Brno E-mail: [email protected] Abstract

Fig. 1: The wiring diagram of the measuring system

This paper describes the design and construction of an electronic interface for the differential capacitance sensor of pressure. The 14-bit precision of measurement is required. The total power consumption below 5 mW and galvanic separation is needed.

The measuring system is supplied over the communication interface of the 4 – 20 mA current loop. The switched mode power supply is used to feed the appropriate low voltage to the microprocessor and also to achieve the galvanic-separated supply for measuring electronics.

1. Introduction The differential measuring system is frequently used for measuring of pressure. The output of the differential pressure sensor is a pair of capacitors. The value of these capacitors can be up to tens of pF. The most important issue is the precision of measuring, therefore the 14-bit AD converter (ADC) is commonly used. The high accuracy ADC is needed to ensure precision in the order of fF. As these values are very low, the measurement is very difficult. The aim of this paper is the describe of the design of low-power and high-precision measuring system for very low capacitors.

3. Differential output of the pressure sensor The output of the differential pressure sensor is a couple of variable capacitors. The dependence of output capacity on differential pressure is shown in Fig. 2. The lay-out of the differential capacitive pressure sensor leads to self compensation of some errors by parasitic capacities, leakage currents and charge injection. 10 C [pF] 8 6 4

2. System topology

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The wiring diagram of the designed measuring system can be observed in Fig. 1. It shows two parts of the electronics of the measuring system. The first part including the main capacitance sensor with related measuring electronics and a A/D converter are galvanicseparated from the circuitry of the current loop. The second part contains the embedded microprocessor, a temperature sensor and circuitry for current loop servicing.

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Fig. 2: Output of the differential pressure sensor

4. Measuring system The measuring system is based on the charge division technique at the capacitors and subsequent measurement of steady voltage on a parallel combination of capacitors (in Fig. 3). The above mentioned measuring technique needs the lowest power consumption as compared with other techniques. The main difficulty of measuring by the charge division technique is the very low capacity of output capacitors. The low value (50 to 250 pF) leads to spontaneous losses of charge. The measurement is significantly affected by many factors. The crucial factor is a parasitic leakage current of analog switches, a charge

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injection from/to analog switches, noise and temperature drift.

Fig. 3: Simplified measuring system

On the basis of the known properties of the real parts such as analog switches and operational amplifiers the equivalent circuit diagram was created (in Fig. 4). The equivalent circuit diagram considers the head parasitic properties that cannot be eliminated.

The measuring process can be separated into two steps – establishing initial conditions and charge division with sampling voltage. Switches S1 and S3 in Fig. 2 are turned on at time t1. The normal capacitor C1 is connected to the reference voltage through the analog switch and charged up to its voltage. At the same time the measured capacitor C2 is connected through the analog switch S3 to the ground and the whole charge accumulated on capacitor C2 is led off. At the time t2 the analog switches S1 and S3 are switched off and switch S2 is switched on. Thereby the charge transport from the normal capacitor C1 to the measured capacitor is allowed. After a sufficiently long time, approximately 10 τ, the voltage on the capacitors is almost stable. At this moment the voltage is sampled by track and hold circuitry. The voltage is held for the following processing by ADC.

6. The reference capacitor The high quality dielectric of the reference capacitor has to be used. There should be a minimal parasitic leakage current and minimal changes of properties depending on temperature. A very important parameter is the absolute value of the capacity of the reference capacitor. It comes out from equation Fig. 4: Equivalent circuit diagram of the measuring system

Great deal of simulation was made with the equivalent circuit diagram. The transient simulation is the key for the measuring system and can be observed in Fig. 5. Figure 5 shows measuring circuitry behavior at point A in Fig.4.

ΔU =

U1 − C R U − CR . − 1 C R + Cmax C R + Cmin

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For the range of the measured capacities between 50 pF and 250 pF and the reference voltage U1 = 3 V the widest range of the final stable voltage is obtained when the reference capacitor is Cr = 110 pF. Fig. 6 shows the dependence of voltage variation on reference capacitance. For Cr = 110 pF the final voltage range is ΔU = 1,1458 V. U [V]

1,20 1,10 1,00 0,90 0,80 0,70 0,60 0,50 0,40 0,30 C [pF]

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Fig. 5: Transient characteristic of measuring system

5. The measurement

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Fig. 6: Dependence of the output voltage variation on the absolute value of normal capacity

The voltage step of LSB is approximately ULSB = 70 μV for the desired accuracy of 14-bits. Therefore, the


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maximal uncorrectable error of the measured voltage must not exceed the value of 35 μV.

7. Measuring errors Many various errors rise during the measuring process. Some of these errors are systematic and can be eliminated or corrected, and some are the uncorrectable accidental errors. The sum of uncorrectable errors must not exceed 35 μV. The analog switches are responsible for highest level of uncertainty in the proposed measuring system shown in Fig. 7. There is more than one reason. The first one is a charge injection from the analog switch into measuring circuitry. During the switch-on phase the charge is moved to switch while during the switch-off phase the charge is removed from switch. From [1] the maximal value of injected charge is approximately 4 pC. The value of this charge leads to change of the measured voltage by about 20mV on the parallel combination of the reference capacitor and the measured capacitor. It is nearly 286 times more than desired value.

Fig. 7: Proposed measuring circuitry

Another problem with the real switches here is the fact that the value of the injected charge hardly depends on the value of voltage on the switch. Thanks to the circuitry shown in Fig. 7 the charge removing from the analog switch which goes switch-off is moved to the analog switch which goes switch-on. It leads to a partial compensation of the injected charge. The value of the injected charge mainly depends on geometric dimensions of the analog switch.

We can suppose that the values of the injected charge of the two equal analog switches in the same conditions should be equal, too. Therefore, the measuring error by injected charge can be effectively corrected. Another error produced by the analog switch is a parasitic parallel input capacity. The capacity error is several units of pF and it may be assumed that the capacity error is, in a short time interval, time independent. This capacity leads to generation of a certain offset on the measured capacitor. One last important source of the measuring error is the leakage current of the analog switch between inputs and ground in switch-on stage (max. 2 nA) and in switch-off stage (max. 1 nA). Even if it’s a relatively well-known short time independent variable; its existence leads to gradual discharging of the parallel combination of the measured and reference capacitors. The conversion time of the high resolution and low power ADCs is usually from tens to hundreds of milliseconds. The voltage downswing on parallel combination of capacitors Cr and Cx is so striking that the result of AD conversion would be worthless. The Track and Hold circuitry is used to maintain the value of the measured voltage during measuring process. This voltage must not fall below the already shown ULSB = 35 μV. Because the T-H circuitry is used behind the input buffer it is possible to use large memory capacitors (order of tens of μF).

8. Track and Hold circuitry The input buffer for T-H circuitry is needed otherwise the measured voltage would be negatively influenced. In the same way, the buffer’s input parasitic capacity and input current would be as small as possible. By using only one buffer for measured voltages separation some additional errors from real buffer are perfectly suppressed. In some cases other errors depending on buffer are also suppressed. There are a couple of analog switches on the output of the buffer. The switches apply output of buffer to the memory capacities. The T-H circuitry is shown in Fig. 9.

Fig. 8: Dependence of injected charge on led voltage [1] Fig. 9: Track and Hold circuitry with input buffer


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Even here the feature of the differential circuitry is used. The possible voltage drop would be equal for both of memory capacities and voltage drops eliminates themselves. There are relatively big capacitors used like memory capacitors (orders of microfarads). Therefore, held voltage is affected by charge injection minimally. However relatively big capacities require long charging time. The main condition for a proper function of the measuring system is very short sampling time. In this place the digital part of electronics is helpful. The microprocessor ensures series of successive samplings to recharging memory capacitor to the exact value of the measured voltage. The possible timeline of memory capacitor recharging is shown in Fig. 10. The sampling timeline would be operationally changed depending on measured voltage.

Fig. 10: Possible timeline of operations

9. Digital part of electronics The present trend is to locate nonvolatile memory containing information about sensor and calibration data directly to the sensor chassis. Thanks to this lay-out the used sensor is uniquely identifiable. The calibrating data allows calibrate measured data processing electronics. In the light of flexibility and consequentiality of the designed device it seems to be the best solution accommodate measuring electronics directly to the sensor. On the basis of the calibrating data correction of the measured values can be calculated. An optimal solution for the measured data processing electronics is the low power microprocessor. The used microprocessor can computes mathematical operations which are needed for the final calibration and its large memory EEPROM or FLASH [2] would be used as a storage of calibration data.

9.1. Calibration of measured values There are many ways how to calibrate the measured values. The easiest way is correction of the measured values using correction table. This is a relatively fast way of correcting nonlinear behavior but it’s not applicable for calibration of high resolution converters above 12bits.

The next fast correction is linear regression, but it can be used only in case of almost linear behavior. An equivalent of the pure linear regression is partial linear regression. The corrected characteristic should be divided to the number of parts and each part has its own function. The number of parts depends only on behavior of corrected characteristic, therefore individual parts would have not same length. Obviously, the most sophisticated solution is an approximation by n-order multinomial. It is possible to linearize the whole conversion characteristic with variable reliability till R = 99,995 %. This feature is balanced by very difficult computing decimal numbers and power functions.

9.2. Designed calibration algorithm As noted previously, a high-precision calibration is desired as well as a minimal power consumption of the used microprocessor. The combination of partial linear regression and the 48-bit extended measured value was chosen. The principle of the extended measured value is simple. On the 32-bit range, the maximal number is already nearly 4,29.109, whereas using the 32-bit number extends 16-bit measured value to virtual 4 decimal points. Due to this extension the high precision of computing is achieved along with relatively fast elaboration. At the first step of elaboration the measured 16-bit number is multiplied by decimal number 104. After that the regression function is applied. At the end the calibrated number is divided by 104 and rounded.

10. Measure of temperature An absolute majority of materials physical properties depend on temperature. That is the reason why an actual value of temperature is needed. The temperature is measured with 12-bit accuracy on the military range –40 to 125°C. In order to measured temperature would be significant, the temperature sensor should be placed as close as possible to the measured capacitors. The silicon paste should be used for the increasing heat flow.

11. Power supply One of the common requirements for the sensors electronics is the galvanic separation from another measuring system and automation, respectively 4-20mA current loop. Practically only one useful solution are switched changers. The total power limitation of 5 mW is needed. Even if the switched changers efficiency reaches up to 96 %, this tiny power demands almost half of the total power for the own function. Available power is only


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2,5mW. This restriction eliminates using competitions circuits [5].

[1] Texas Instruments, “Datasheet of MAX4595”, 10.10.2005 http://www-s.ti.com/sc/ds/max4595.pdf

12. Conclusion

[2] Atmel Corporation, “Datasheet of ATMEGA88”, 10.10.2005, http://www.atmel.com/dyn/resources/prod_documents/do c2545.pdf

On the basis of many simulations and measuring the proposed circuitry can be used for the precision measuring of the differential capacitance pressure sensor. The 14-bit resolution of the measured values of capacities can be obtained if the main conditions are complied.

13. Acknowledgements The research has been supported by Czech Ministry of Industry within the frame of the Research Plan MSM0021630503 MICROSYN, by the Czech Grant Agency as the project GA102/03/0619 Smart Microsensors and Microsystems for Measurement and Regulation and projects GA102/03/H105, FF-P/112 and FT-TA/050.

14. References

[3] Atmel Corporation, “Preserve energy with Low 10.10.2005, Power AVR Microcontrollers”, http://www.atmel.com/dyn/resources/prod_documents/do c4060.pdf [4] Atmel Corporation, “Enhancing ADC resolution by over sampling”, 10.10.2005, http://www.atmel.com/dyn/resources/prod_documents/do c8003.pdf [5] Michal Brychta, “Measure Capacitive Sensors With A Sigma-Delta Modulator”, ED Online, April 28, 2005, http://www.elecdesign.com/Articles/Index.cfm?ArticleID =10185&pg=


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