Flexible Power Electronic Transformer - IEEE Xplore

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 25, NO. 8, AUGUST 2010

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Flexible Power Electronic Transformer Mehran Sabahi, Member, IEEE, Ali Yazdanpanah Goharrizi, Student Member, IEEE, Seyed Hossein Hosseini, Member, IEEE, Mohammad Bagher Bana Sharifian, and Gevorg B. Gharehpetian, Senior Member, IEEE

Abstract—This paper proposes a new modular flexible power electronic transformer (FPET). The proposed FPET is flexible enough to meet future needs of power electronic centralized systems. The main feature of the FPET is the independent operation of modules each of which contains one port. Each port can be considered as input or output, because bidirectional power flow is provided. The modules are connected to a common dc link that facilitates energy transfer among modules as well as ports. Therefore, a multiport system is developed, which the ports can operate independently. This merit is important for applications, where input and output voltages are different in many parameters. A comparison study is carried out to clarify the pros and cons of expandable FPET. In addition, the measurement results of a laboratory prototype are presented to verify the capabilities of FPET in providing different output waveforms and controlling load side reactive power. Index Terms—DC link, flexible power electronic transformer (FPET), high-frequency isolation transformer, pulsewidth modulation (PWM).

I. INTRODUCTION OWER electronic transformers (PETs) are proposed to replace conventional transformers and perform voltage regulation and power exchange between generation and consumption by electrical conversion [1]–[5]. The previous researches show that PETs have a great capacity to receive much more attention due to their merits such as high-frequency link transformation and flexible regulation of the voltage and power. Although many studies have been conducted on application and control of PET in power systems [1]–[8], less attention is paid to the areas of the circuit topologies [7] and [8]. The topology of PET can be developed in such a way to achieve multiport electrical system that converts variable input waveform to the desired output waveform. In addition, for higher voltage applications or three phase systems, the topology is expandable as it is modular. In this paper, a new PET topology named flexible power electronic transformer (FPET) is proposed. As shown in Fig. 1, it is constructed based on modules and a common dc link, which

P

Manuscript received June 29, 2009; revised September 26, 2009 and November 30, 2009; accepted January 1, 2010. Date of current version July 16, 2010. Recommended for publication by Associate Editor F. Wang. M. Sabahi, S. H. Hosseini, and M. B. B. Sharifian are with the Faculty of Electrical and Computer Engineering, University of Tabriz, Tabriz 51666-16471, Iran (e-mail: [email protected]; [email protected]; [email protected]). A. Y. Goharrizi is with the Faculty of Electrical and Computer Engineering, University of Tabriz, Tabriz 51666-16471, Iran, and also with the Islamic Azad University of Sofian, Sofian 53861, Iran (e-mail: [email protected]). G. B. Gharehpetian is with the Electrical Engineering Department, Amirkabir University of Technology, Tehran 15914, Iran (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TPEL.2010.2040840

Fig. 1.

Main concept of proposed FPET.

is used to transfer energy between ports and isolate all ports from each other. In this bidirectional topology, each port can be considered as an input or output. Each module consists of three main parts, including modulator, demodulator, and highfrequency isolation transformer (HFIT). The modulator is a dc– ac converter and the demodulator is an ac–ac converter; both with bidirectional power flow capability. Each module operates independently and can transfer power between ports. These ports can have many different characteristics, such as voltage level, frequency, phase angle, and waveform. As a result, FPET can satisfy almost any kind of application, which are desired in power electronic conversion systems and meet future needs of electricity networks. Considering this point, it is named flexible. The simulation results of high-voltage application are given to clarify the advantages of the proposed FPET over the recently developed PETs [9]. To show the flexibility of the proposed PET, a prototype is built and tested. II. PROPOSED POWER CIRCUIT OF FPET The proposed circuit is shown in Fig. 2. It should be mentioned that the proposed topology can be expanded by connecting modules in series or parallel to obtain higher voltage or current ratings, and to form star/delta connections for three phase applications. As shown in Fig. 2(a), each port is composed of a fullbridge dc-link inverter (FBDCI), HFIT, and a cycloconverter. This topology consists of independent and similar modules and each port can work independently. Thus, the analysis of one port is sufficient to introduce whole topology. The FBDCI (modulator) can operate as an inverter when it converts the dc-link voltage to an ac waveform at the HFIT side. It can operate as an active rectifier when it converts the ac waveform of the HFIT to the dc-link voltage. The FBDCI is used to achieve zero-voltage level, adjustable pulsewidth, and symmetrical switching. In addition, the number of switches can be reduced to obtain simpler

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Fig. 2.


Proposed circuit of the FPET. (a) Basic topology and (b) reduced switch topology.

circuit than the latter, shown in Fig. 2(b). In this case, one of the half-bridge circuits can be considered as the reference or master leg. Once gate pulses for the master leg (i.e., switches and ) are provided, the gate pulses of the other legs (slave legs) have a phase shift respect to the master leg. Using this control strategy, the number of switches can be reduced to half. The modulator can be described as follows: 1) bidirectional power flow capability; 2) adjustable switching frequency that feet voltage pulses frequency into the passband of HFIT; and 3) Stored energy in the dc link (if the modulator is in active rectifier mode). For cycloconverters, several circuit topologies can be proposed using unidirectional or bidirectional switches [10]–[12]. In this paper, a typical cycloconverter with two bidirectional switches operates as the demodulator. The demodulator converts high frequency voltage (i.e., ) to low frequency voltage (i.e., Vpr 1 ) and vice versa. The specifications of the demodulator are listed as follows: 1) bidirectional power flow capability; and 2) Providing zero voltage switching by turning the switches of cycloconverter ON/OFF, while voltage of HFIT riches to zero. III. MODULATION AND DEMODULATION OPERATION PRINCIPLES The well-known phase shift modulation (PSM) method is shown in Fig. 3. The definition of parameters is given in Table I.

Fig. 3.

Principle of PSM method.

The voltage regulation is performed by the FBDCI using PSM method. The cycloconverter chooses the PSM pulses in such a way to provide positive or negative voltage polarity at the output. In this figure, the cycloconverter provides positive output voltage polarity as an example. On one hand, the switches of cycloconverter turn ON/OFF with a time delay (Tcd ) respect to those of FBDCI, so they operate under zero voltage condition. On the other hand, the switches have a small overlapping time to provide a path for Lf current to avoid high stresses at switching instants. Thus, the switches operate at soft switching condition. The leakage inductance of HFIT should be minimized as much

SABAHI et al.: FLEXIBLE POWER ELECTRONIC TRANSFORMER

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TABLE I DEFINITION OF PARAMETERS

as possible. In practice, snubber circuits must be used to damp the stored energy in the leakage inductance of HFIT. According to Fig. 3, the duty cycle of FBDCI is defined as follows: D(kTs ) =

2Ton (kTs ) , Ts

k = 1, 2, 3, ....

(1)

The modulated voltage at the secondary side for one duty cycle is expressed by (2) Vs = N V p .

(2)

The modulated voltage at the output of cycloconverter (Vc ) is determined as follows: Vc (t) = sign(t) |N Vp (t)| = sign(t)N Vd (t), sign(t) = 1 or − 1, (k − 1)Ts < t < kTs ,

k = 1, 2, 3, ...

Fig. 4.

Schematic presentation of PSM controller.

between the port and the grid. So, a controllable voltage at the output of cycloconverter can be obtained and it is given by vc i (t) = vRef i (t)

where vRef i (t) is the reference voltage. According to (4), one may obtain the following approximation: vRef i (t) ≈ Kc sign∗i ((k + 1)Ts )Ni Nd (kTs )Di∗ ((k + 1)Ts ),

(3) where sign(tk ) function determines the polarity of Vc that can be positive or negative according to the desired output voltage and presented by (4), as shown at the bottom of the page. A. PSM Control Circuit The control circuit is responsible for providing pulse gate of dc link switches and the cycloconverter. The implementation of PSM is shown in Fig. 4. The input data address consists of four lines. The first line is polarity of output voltage signi . The second line is switch-enabled of cycloconverter (EnableC i ). The third line is switch-enabled of dc link (EnableS i ). The fourth line provides the duty cycle data of the ith port. The enabled lines are provided by the startup and protection circuits.

(5)

kTs < 1 < (k + 1)Ts

(6)

where the asterisk symbols show the next stage values. Therefore, the duty cycle and the sign function are achieved as follows: |vRef i ((k + 1)Ts )| ∗ Di ((k + 1)Ts ) ≈ , Kc Ni Vd (kTs ) 0< D < 1 sign∗i ((k + 1)Ts ) = sign[vRef i ((k + 1)Ts )]. (7) Because of high switching frequency, it is expected to assume vRef i is constant over time period of kTs < t < (k + 1)Ts . The duty cycle is a function of dc-link voltage (Vd (kTs)) and the turn winding of the HFIT at the ith port. The block diagram of controller is shown in Fig. 5.

B. Utilization of Ports as a Voltage Source As an example when a port (assuming ith port) is designed to operate as a voltage source, it can provide a constant voltage regardless of the active or reactive power that is exchange

IV. ENERGY BALANCE IN FPET In every system, there is a balance among losses, input energy and output energy. This balance for FPET is presented as

  1 ⇒ Ga (t) = G1 (t − Tcd ) and Gb (t) = G2 (t − Tcd ) sign(t) =



, −1 ⇒ Gb (t) = G1 (t − Tcd ) and Ga (t) = G2 (t − Tcd )

(k − 1)Ts < t < kTs , k = 1, 2, 3, ....

(4)

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Fig. 5.


Control circuit of a typical port that operates as a voltage source. TABLE II DESCRIPTION OF PARAMETERS PREARRANGED IN (10)

follows: n

Wi + WC d + Wloss = 0

(8)

i=1

where Wi , WC d , and Wloss are the input/output energy, stored energy at dc link and losses, respectively. Neglecting the power losses, (8) can be approximated by n ∆ Pi ≈ −∆PC d . (9)

Fig. 6.

Simplified diagram of FPET.

i=1

To achieve power equilibrium in Cd and have constant dclink voltage, some of the ports should absorb and inject desired active power. The algorithm for regulation of dc-link voltage is as follows: Step 1: At the start-up instant, following two methods can be used to charge the dc-link capacitor to the desired value. 1) The dc-link capacitor can be charged by an extra dc source. As the desired dc-link voltage achieved, the dc source should be disconnected. 2) The cycloconverter can provide a high frequency voltage across HFIT. When the voltage passes through HFIT, it changes to a dc voltage across dc-link capacitor by the body diodes of FBDCI switches. The dc voltage can charge the capacitor considering the winding ratio of HFIT. The startup current is limited by Lf . Step 2: dc-link voltage checking. 1) If Vd,Ref − ∆Vd,Ref < v d (t) < Vd,Ref + ∆Vd,Ref , then there is no need for adjustments. The ∆Vd,Ref is a fraction of Vd,Ref that is required to provide Hystersis band. 2) If Vd,Ref − ∆Vd,Ref > v d (t) or v d (t) > Vd,Ref + ∆Vd,Ref , then voltage should be regulated and the port powers should be adjusted. Step 3: Return to the second step. A. Balancing Ports For another solution to regulate voltage of dc link, some ports are considered as “balancing ports” that provide energy to balance dc-link voltage in FPET. One of the main objectives

of these kinds of ports is to control voltage level in the dc-link voltage, particularly when over voltage or voltage drop occurs in the dc link. Assuming the ith port is chosen as the balancing port, the main component of the cycloconverter voltage, and output of the port are given as follows: √ vc i (t) = 2Vc i sin(2πfi t − φc i ) → Vc i φc i √ , vpr i (t) = 2Vpr i sin(2πfi t − φp i ) → Vpr i φp i δ i = φc i − φ p i .

(10)

The definition of the parameters is given in the Table II. Therefore, neglecting the resistance of output filter inductance, the active power of the port is obtained as follows: Pi =

Vc i Vpr i sin δi . 2πfi Lf

(11)

Applying the differences between Vd and Vd,Ref as an error signal to a typical PI controller, the value of required Pi can be estimated. According to (6) and (7), the duty cycles are achieved. V. DESIGN PROCEDURE A. DC-Link Capacitor Fig. 6 shows the voltage and currents of all ports and the dc link capacitor. The following equation presents the instantaneous power balance of the losses in FPET. n i=1

ipr i (t)vpr i (t) + vd (t)id (t) = 0.

(12)


Fig. 7.

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Proposed HV FPET.

The voltage and current of ports can have different polarity and directions. If the currents and voltages of ports have sinusoidal waveforms, then (12) can be rewritten as follows: n

∆Vd 1 = Im Vm cos(2ωi t − φi − θi ). P˜ ≈ Vd,Ref Cd ∆t 2 i=1 i i n

(16) Im i sin(ωi t − φi )Vm i sin(ωi t − θi )+vd (t)id (t) = 0.

i=1

(13) Now, the input power of dc link can be expressed as follows:

∆Vd ≈

1 Im Vm cos(2ωi t − φi − θi ) vd (t)id (t) = 2 i=1 i i n

n

(14) This input power consists of two components. The first component is the pulsation power (P˜ ) with angular frequency of 2ω i and the second one is the dc power (P¯ ). Assuming Vd,Ref as the voltage of capacitor and Id as the average current, (14) can be rewritten as follows: n i=1

Im i Vm i cos(φi − θi )

n 1 Pi 1 . Cd Vd,Ref i=1 ωi

(17)

Thus, the minimum value of Cd can be calculated for the maximum voltage ripple.

1 Im Vm cos(φi − θi ) = P˜ + P¯ . − 2 i=1 i i

1 P¯ ≈ Vd,Ref Id = − 2

The ripple voltage of the dc-link capacitor (∆Vd ) can be approximated as follows:

(15)

B. Reference Voltage of DC Link and Winding Ratio of HFIT From practical point of view, lower dc-link voltage results in lower voltage stress of switches. But according to (17), as Vd,Ref decreases, the voltage ripple increases. In addition, the decrease of the dc-link voltage increases the current of dc link switches. Consequently, by selecting an appropriate dc link reference voltage (Vd,Ref ) and the maximum ripple voltage, the minimum dc-link voltage (Vd, m in ) can be determined. In the worst condition, the lowest dc-link voltage (Vd,m in ), maximum duty cycle (D = Dm ax ) and the maximum magnitude of desired

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TABLE III PARAMETERS OF PETS

Fig. 8.

Fig. 9.

Port voltage and current of HV FPET.

voltage (Vi,m ax ) can determine the winding ratio as follows: Vi,m ax Ni > . Kc Vd,m in Dm ax

(18)

Load voltage and current of three-phase output.

by ∆if i ≈ (Ni Vd,m ax + Vi,m ax ) Ts 1 . Lf > (Ni Vd,m ax + Vi,m ax ) fs Ii,m ax

Lf i

(20) (21)

C. Matching Inductance Lf Matching inductance Lf should limit the output current to its maximum acceptable value (Ii,m ax ) during the switching period (Ts ). For the ith port, the following assumptions can be considered:  Vs = Ni Vd,m ax      ∆if < Ii,m ax i (19)  Vpr = −Vi,m ax     Rf i ≈ 0 where ∆if i is the variation of the cycloconverter current for one switching period. Based on these assumptions, Lf is determined

VI. APPLICATIONS The proposed FPET is flexible enough to be used in high voltage (HV) and low voltage (LV) applications. In this section, two main studies are presented for both applications. A. High Voltage Applications In order to provide a HV application, the modules of PETs are connected in series [9], [13], and [14]. The cascaded H-bridge multilevel PET has been proposed in [9]. The advantages of this PET are: the low switching frequency, the low input current harmonics, the power factor correction, and the reduction of


Fig. 10.

Two-port FPET circuit diagram.

Fig. 11.

Laboratory prototype.

the input voltage distortion at the output side. Fig. 7 shows the proposed HV FPET, which should be compared with the PET, suggested in [9]. As can be seen in this figure, the ports one to five, i.e., P1 , P2 ,. . ., P4 are connected in series to increase the rating of the input voltage. The RC circuits (Rs and Cs ) are connected to each port to divide high input voltage equally among the ports. The sixth, seventh, and eight ports are connected to a low voltage three-phase load. Table III lists the parameters of both FPET and cascaded H-bridge multilevel PET. Fig. 8 shows the voltage and the current of one of the five ports of HV FPET. Considering the phase of the sinusoidal current

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TABLE IV PARAMETERS OF PROTOTYPE

waveform, the port draws power from the utility grid (v1 , see Fig. 7) with almost unity power factor. Fig. 9 shows the three phase balanced load voltages and currents.

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Fig. 12.


Voltage and current of ports. (a) P1 and (b) P2 .

Fig. 14.

Voltage and current waveforms of switches (a) Sa , (b) S1 , and (c) S3 .

On the other hand, FPET has the capability of the bidirectional power flow, while the multilevel PET is unidirectional. It must be mentioned that, FPET has one dc link and one dc capacitor but multilevel PET has two dc links in each module. In addition, the output ports of FPET can be connected in star configuration to provide a three phase four-wire system with independent phase voltage control. Fig. 13.

Voltage and current of P1 . (a) Leading current and (b) lagging current.

B. Low-Voltage Power Electronic Application In order to study the capability of FPET to reduce the input voltage disturbances such as voltage swell and sag, 50% voltage swell and 50% voltage sag is applied to the supply of FPET. Fig. 10 shows the input, load, and dc-link voltages. This is clear that the output voltage, i.e., port 6 remains almost constant during voltage sag and swell, respectively. These simulations show that the multilevel PET proposed in [9] and FPET have the same capability of the power factor correction and power quality enhancement. The advantage of multilevel PET over FPET is its lower harmonic components in the input current.

In power electronic applications, the voltage conversion and reactive power control are regarded as the center of interest. The main goal of this study is to outline the capability of the FPET to provide both the desired output waveform and the input reactive power control. The two-port FPET circuit diagram and its implemented prototype are shown in Figs. 10 and 11, respectively. The prototype consists of two ports. As shown in Fig. 10, the first port (P1 ) is modeled by L1 and v1 and connected to the utility grid. The second one (P2 ) is connected to the inductive load. Table IV lists the parameters of


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TABLE V COMPARISON STUDY OF FPET AND THOSE PROPOSED IN [4] AND [12]

the prototype. For the first stage, a 50-Hz waveform should be converted to 60-Hz sinusoidal waveform by this proposed circuit. The measured and simulated waveforms of the voltages and currents of ports, P1 and P2 are shown in Fig. 12. The voltage and current of P1 are in phase and the frequency of the output voltage is 60 Hz. As a result, it can be deduced that the FPET has the capability of power factor correction and regulation of the output frequency. As additional example, two reactive power operation conditions with leading and lagging current are shown in Fig. 13. The measured voltage and current of switch of cycloconverter (i.e., Sa ) and the switches of FBDCI (i.e., S1 and S3 , respectively) are shown in Fig. 14. As it is clear from Fig. 14(a) that the switch Sa experiences zero-voltage condition while it turns on or turns OFF. The switches S1 and S3 turn OFF while current is zero. VII. COMPARISON STUDY A comparison study is given to clarify the advantageous and disadvantageous of the FPET. A three-phase system, contains six ports, is compared to the similar PETs. First, some of the pros and cons of bidirectional FPET in comparison to the unidirectional topologies should discuss. In the unidirectional systems, input power factor is not controllable but in bidirectional structures input or even output power factor can be adjusted. This means that the reactive and active power of each port can be regulated. Also for DG systems like wind turbine, bidirectional

capability is indispensable [12]. Energy management for energy efficient systems is another application of this feature [11]. A detail comparison study (e.g., cost, efficiency, quality, etc.) is given in Table V to clarify the pros and cons of FPET and the existing topologies proposed in [4] and [12]. As can be seen from Table V, conversion efficiency of FPET is relatively low in comparison to the similar circuits topologies proposed in [4] and [12]. The main reason is the usage of power snubber, and voltage clamp circuits, which damp absorbed energy in leakage inductors of HFIT. To reduce the size of protection circuits in FPET, a PSM approach is utilized, so the cycloconverter switches just select the PSM pulses and can commutate naturally. Therefore, the switches communicate at almost zero voltage. In addition, because of overlap technique the voltage surge is reduced over the switches and the continuous current flow in the output filter (Lf ) is not interrupted. In addition, Table V shows some of the most noticeable applications of FPET. Dynamic voltage restorer (DVR) [15] and active filter (AF) [16] applications can be satisfied by the FPET, because it can connect to the grid in series or/and in parallel. Desired voltage and current can provide by the flexibility of FPET in providing various waveforms (see Section VI). FPET can provide desired waveform in each phase (or port) independently, so this can be used in universal power quality conditioner (UPQC) [17]. FPET can transfer active and reactive power from one port or phase to another port or one phase. This in power distribution system is very useful for interline power flow controller (IPFC) [18]. Additionally, FPET can provide symmetrical three-phase voltage from an asymmetrical ac source in the form

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of an uninterruptible power supply application (UPS). FPET can play a role in providing useful power from variable low-voltage dc sources. That is suitable for renewable energy applications such as photovoltaic and fuel cell [11]. Design simplicity and expandability (to achieve higher ratings) are other advantageous of FPET.

[12] H. J. Cha and P. N. Enjeti, “A three-phase AC/AC high-frequency link matrix converter for VSCF applications,” in Proc. IEEE 34th Annu. Conf. (PESC 2003), Jun., vol. 4, pp. 1971–1976. [13] J. S. Lai, A. Maitra, A. Mansoor, and F. Goodman, “Multilevel intelligent universal transformer for medium voltage applications,” in Proc. IEEE Ind. Appl. Conf., Oct. 2005, vol. 3, pp. 1893–1899. [14] H. Iman-Eini and S. Farhangi, “Analysis and design of power electronic transformer for medium voltage levels,” in Proc. IEEE Power Electron. Spec. Conf., Jun. 2006, pp. 1–5. [15] P. Roncero-Sánchez and E. Acha, “Dynamic voltage restorer based on flying capacitor multilevel converters operated by repetitive control,” IEEE Trans. Power Del., vol. 24, no. 2, pp. 951–960, Apr. 2009. [16] S. Chakraborty and M. G. Simões, “Experimental evaluation of active filtering in a single-phase high-frequency AC microgrid,” IEEE Trans. Energy Convers., vol. 24, no. 3, pp. 673–682, Sep. 2009. [17] J. A. Muñoz, J. R. Espinoza, L. A. Moran, and C. R. Baier, “Design of a modular UPQC configuration integrating a components economical analysis,” IEEE Trans. Power Del., vol. 24, no. 4, pp. 1763–1772, Oct. 2009. [18] S. Bhowmick, B. Das, and N. Kumar, “An advanced IPFC model to reuse Newton power flow codes,” IEEE Trans. Power Syst., vol. 24, no. 2, pp. 525–532, May 2009.

VIII. CONCLUSION Based on the requirement of a flexible power conversion system, FPET is proposed to facilitate many requirements that are expected in power electronic and distribution systems. The proposed topology is flexible enough to provide bidirectional power flow and has as many ports as it is required. For low-voltage application, FPET can correct power factor and can adjust the waveform and frequency of the output voltage. The proposed topology can be expanded for high voltage and high current applications. The dc link plays a significant role to provide energy balance, power management in the circuit and independent operation of ports. The measurement results verify the basic theoretical concepts of this paper. The advantages of the FPET are: bidirectional power flow capability of ports, module-based topology, which can be used in different forms, independent operation of ports, flexibility in power amount and direction in all ports, and double galvanic isolation between each port, as well as using only one storage element. REFERENCES [1] S. H. Hosseini, M. B. Sharifian, M. Sabahi, A. Yazdanpanah, and G. H. Gharehpetian, “Bidirectional power electronic transformer for induction heating systems,” in Proc. Can. Conf. Electr. Comput. Eng., May 4–7, 2008, pp. 347–350. [2] S. H. Hosseini, M. Sabahi, and A. Y. Goharrizi, “Multi-function zerovoltage and zero-current switching phase shift modulation converter using a cycloconverter with bidirectional switches,” IET Power Electron. JNL, vol. 1, no. 2, pp. 275–286, Jun. 2008. [3] M. Sabahi, S. H. Hosseini, M. B. Sharifian, A. Yazdanpanah, and G. H. Gharehpetian, “A three-phase dimmable lighting system using a bidirectional power electronic transformer,” IEEE Trans. Power Electron., vol. 24, no. 3, pp. 830–837, Mar. 2009. [4] D. Wang, M. Chengxiong, L. Jiming, S. Fan, and C. Luonan, “The research on characteristics of electronic power transformer for distribution system,” in Proc. IEEE Transmiss. Distrib. Conf. Exhib. Asia Pacific, 2005, pp. 1–5. [5] M. Huasheng, Z. Bo, Z. Jianchao, and L. Xuechao, “Dynamic characteristics analysis and instantaneous value control design for buck-type power electronic transformer (PET),” in Proc. IEEE Annu. Conf. Ind. Electron. Soc. IECON, Nov. 2005, pp. 1043–1047. [6] H. Wrede, V. Staudt, and A. Steimel, “Design of an electronic power transformer,” in Proc. IEEE 28th Annu. Conf. Ind. Electron. Soc., 2002, vol. 2, pp. 1380–1385. [7] J. Aijuan, L. Hangtian, and L. Shaolong, “A new high-frequency AC link three-phase four-wire power electronic transformer,” in Proc. IEEE Conf. Ind. Electron. Appl., May 2006, pp. 1–6. [8] H. Krishnaswami and V. Ramanarayanan, “Control of high-frequency AC link electronic transformer,” IEE Proc. Elect. Power Appl., May 2005, vol. 152, no 3, pp. 509–516. [9] S. Farhangi, H. Iman-Eini, J. L. Schanen, and J. Aime, “Design of power electronic transformer based on cascaded H-bridge multilevel converter,” in Proc. IEEE Int. Symp. Ind. Electron., Jun. 2007, pp. 877–882. [10] D. Chen and J. Liu, “The uni-polarity phase-shifted controlled voltage mode AC-AC converters with high frequency AC link,” IEEE Trans. Power Electron., vol. 21, no. 4, pp. 899–905, Jul. 2006. [11] H. Krishnaswami and N. Mohan, “Three-port series-resonant DC–DC converter to interface renewable energy sources with bidirectional load and energy storage ports,” IEEE Trans. Power Electron., vol. 24, no. 10, pp. 2289–2297, Oct. 2009.

Mehran Sabahi (M’09) was born in Tabriz, Iran, in 1968. He received the B.Sc. degree in electronic engineering from the University of Tabriz, Tabriz, the M.Sc. degree in electrical engineering from Tehran University, Tehran, Iran, and the Ph.D. degree in electrical engineering from the University of Tabriz, in 1991, 1994, and 2009, respectively. In 2004, he joined the Faculty of Electrical and Computer Engineering, University of Tabriz, where he has been an Assistant Professor since 2009. His current research interests include power electronic converters and power electronic transformers.

Ali Yazdanpanah Goharrizi (S’09) received the B.Sc. degree in electrical engineering from Shahid Bahonar University of Kerman, Kerman, Iran, and the M.Sc. degree in power electrical engineering in 2007 from the University of Tabriz, Tabriz, Iran, where he is currently pursuing his Ph.D. through the Faculty of Electrical and Computer Engineering. He is also an academic Member and Lecturer at Islamic Azad University of Sofian, Sofian, Iran. He was honored as exceptional talent at the University of Tabriz, in 2009. His research interests include power electronic application in renewable energy systems, energy management in power electronic systems, modeling and control of power electronic systems, power electronic converters for dimmable lighting and induction heating applications, and power quality enhancement, and flexible AC transmission system (FACTS).


Seyed Hossein Hosseini (M’93) was born in Marand, Iran, in 1953. He received the M.S. degree from the Faculty of Engineering, University of Tabriz, Tabriz, Iran, in 1976, the D.E.A. degree from the Institut National Polytechnique de Lorraine (INPL), Nancy, France, in 1978, and the Ph.D. degree from INPL, in 1981, all in electrical engineering. In 1982, he joined the Faculty of Electrical and Computer Engineering, University of Tabriz, as an Assistant Professor, where from 1990 to 1995, he was an Associate Professor, and since 1995, he has been a Professor. From September 1990 to September 1991, he was a Visiting Professor with the University of Queensland, Brisbane, Qld., Australia, from September 1996 to September 1997, he was a Visiting Professor with the University of Western Ontario, London, ON, Canada. His current research interests include power electronic converters, matrix converters, active and hybrid filters, application of power electronics in renewable energy systems and electrified railway systems, reactive power control, harmonics and power quality compensation systems such as static VAR compensator (SVC), universal power quality conditioner (UPQC), flexible AC transmission system (FACTS) devices.

Mohammad Bagher Bana Sharifian was born in 1965. He received the B.Sc. and M.Sc. degrees in 1989 and 1992, respectively, and the Ph.D. degree in electrical engineering in 2000, all from the University of Tabriz, Tabriz, Iran, where he studied electrical power engineering. In 1992, he joined the Electrical Engineering Department, University of Tabriz, as a Lecturer. In 2000, he rejoined the Electrical Power Department, Faculty of Electrical and Computer Engineering, University of Tabriz, as a Professor, where he is currently an Associate Professor. His research interests include design, modeling and analysis of electrical machines, transformers, and electric and hybrid electric vehicle drives.

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Gevorg B. Gharehpetian (M’00–SM’08) was born in Tehran, Iran, in 1962. He received the B.S. degree with first class Honors from Tabriz University, Tabriz, Iran, in 1987, and the M.S. degree from Amirkabir University of Technology (AUT), Tehran, in 1989, and the Ph.D. degree from Tehran University, Tehran, Iran, in 1996, all in electrical engineering. As a Ph.D. student, he received scholarship from DAAD (German Academic Exchange Service) from 1993 to 1996. He was with the High Voltage Institute, Rheinisch-Westfälische Technische Hochschule (RWTH Aachen), Aachen, Germany.. He is currently at the Electrical Engineering Department, Amirkabir University of Technology, Tehran. He is the author of more than 270 journal and conference papers. His teaching and research interests include power system and transformers transients, flexible AC transmission system (FACTS) devices, and high voltage dc transmission. Dr. Gharehpetian was selected by the Ministry of Higher Education as a distinguished Professor and was the recipient of the National Prize of Iran, in 2008.