Grid-tied Power Converter for Battery Energy Storage ... - KPubS

ISSN(Print) 1975-0102 ISSN(Online) 2093-7423

J Electr Eng Technol Vol. 8, No. 6: 1400-1408, 2013 http://dx.doi.org/10.5370/JEET.2013.8.6.1400

Grid-tied Power Converter for Battery Energy Storage Composed of 2-stage DC-DC Converter Do-Hyun Kim*, Yoon-Seok Lee*, Byung-Moon Han†, Ju-Yong Kim** and Woo-Kyu Chae** Abstract – This paper proposes a new grid-tied power converter for battery energy storage, which is composed of a 2-stage DC-DC converter and a PWM inverter. The 2-stage DC-DC converter is composed of an LLC resonant converter connected in cascade with a 2-quadrant hybrid-switching chopper. The LLC resonant converter operates in constant duty ratio, while the 2-quadrant hybridswitching chopper operates in variable duty ratio for voltage regulation. The operation of proposed system was verified through computer simulations. Based on computer simulations, a hardware prototype was built and tested to confirm the technical feasibility of proposed system. The proposed system could have relatively higher efficiency and smaller size than the existing system. Keywords: Grid-tied power converter, 2-stage DC-DC converter, LLC resonant converter, 2-quadrant hybrid-switching chopper, Battery energy storage.

DC-DC converter with a soft-switching one [13-16]. However, careful attention should be paid to the resonant problem and harmonics generation due to adding the softswitching circuit. This paper proposes a new grid-tied power converter for the battery energy storage, which is composed of a 2-stage DC-DC converter and a PWM inverter. The 2-stage DCDC converter consists of an LLC Half-Bridge resonant converter and a 2-quadrant hybrid-switching chopper. Simulation model was developed to analyze the operation of proposed converter. Based on simulation results, a 10kW hardware prototype was built and tested to confirm the hardware implementation.

1. Introduction The output power of renewable energy source, such as a wind power and Photovoltaic power, varies intermittently depending on weather conditions. Also timing discrepancy occurs between the power generation and the power demand. In order to eliminate this weak point, a battery energy storage, which is composed of power converter and battery, is provided rapidly [1-4]. The power converter for battery energy storage regulates the charging and discharging current, and controls the active and reactive current flowing into or from the grid. The power converter for battery energy storage requires high performance, high efficiency, and small size. It also requires electrical isolation between the AC side and battery side for safety purpose [5-9]. The power converter for battery energy storage can be designed with various topologies according to the power rating, performance, efficiency and safety. The simplest topology can be designed with a PWM converter and 3phase transformer, in which the modulation index of PWM converter is controlled according to the voltage variation of battery. The insertion of DC-DC converter between the PWM converter and the battery can offer wider operation range and more flexible control [10-13]. However, the system efficiency is relatively lower due to the rise of switching loss. In order to reduce the switching loss, many researchers already proposed replacing the hard-switching

2. Proposed Power Converter 2.1. System configuration The power converter for battery energy storage can be simply designed with a PWM converter with 3-phase transformer. This converter has simple structure and high efficiency, but the harmonic level of output current increases when the modulation index of PWM converter is low. So, the operation range of battery voltage is narrow, and the flexibility of current control is small. In order to solve these weak points, a DC-DC converter was inserted between the 3-phase PWM converter and the battery as shown in Fig. 1(a). The PWM converter maintains the DC link voltage constant, while the DC-DC converter controls the charging and discharging current of battery. This converter has wide range of current control which offers independent control in the AC current and the charging and discharging current. Also, harmonic reduction

†

Corresponding Author: Dept. of Electrical and Electronic Engineering, Myongji University, Korea. ([email protected]) * Dept. of Electrical and Electronic Engineering, Myongji University, Korea. ([email protected]) ** KEPCO Research Institute, Smart Energy Lab, Korea. ([email protected]) Received: January 15, 2013; Accepted: June 5, 2013

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Do-Hyun Kim, Yoon-Seok Lee, Byung-Moon Han, Ju-Yong Kim and Woo-Kyu Chae

2.2. LLC Resonant converter The DC-DC converter for charging and discharging the battery requires stable power control, highly efficient power conversion, and reliable power transfer regardless the voltage variation in battery. Also reduction of physical size is a key issue, which can be implemented by the isolated high-frequency transformer. In this paper LLC resonant converter was selected among all the softswitching resonant converters. The LLC resonant converter transfers power with relatively high efficiency through resonant phenomenon. The LLC resonant converter can operate in ZVS mode. It does not require connecting an additional inductance for resonance. And it has narrow variation in operation frequency regardless the load variation. Fig. 3 shows the operation mode of LLC resonant converter. The converter operation is divided into four modes according to the time interval. In mode 1, IGBT switch SaH turns on and IGBT switch SaL turns off as shown in Fig. 3(a). The primary current flows through the transistor in SaH, the magnetizing inductance Lp, and the DC capacitor Cdc. And the secondary current flows through the resonant capacitance Cr, the filter capacitor, the diode in SbL, the leakage inductance Lr and the magnetizing inductance Lp. The source power in primary side is transferred to the secondary side, and the resonant capacitor Cr is charged. In mode 2, both IGBT switches SaH and SaL turn off as shown in Fig. 3(b). The primary current flows through the DC capacitor Cdc, the diode in SaL, the magnetizing inductance Lp. The secondary current flows through the resonant capacitor Cr, the diode in SbL, the leakage inductance Lr, and the magnetizing inductance Ls. No

(a) Non-Isolated hard-switching DC-DC converter

(b) Isolated soft-switching DC-DC converter Fig. 1. Power Converter for Battery Charger is possible. However, the system efficiency is low and the system size is bulky due to inserting one more stage. In order to eliminate these weak points, many researchers proposed replacing the non-isolated hardswitching converter with an isolated soft-switching converter, and replacing the line frequency transformer with the coupling reactors as shown in Fig. 1(b). A typical soft-switching converter is an active clamp converter, which has rather higher switching loss [17]. Fig. 2 shows whole structure of the proposed power converter including the control system for each stage. The PWM inverter controls the active power and the DC link voltage. The 2-quadrant hybrid-switching DC-DC converter operates with variable duty ratio to control the charging and discharging current, while the LLC resonant converter operates with a fixed duty ratio. The proposed DC-DC converter offers higher efficiency with soft switching in LLC resonant converter and hybrid switching in 2-quadrant chopper [18].

Fig. 2. Configuration of Proposed Power Converter

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Grid-tied Power Converter for Battery Energy Storage Composed of 2-stage DC-DC Converter

Fig. 4. Gate pulse for Hybrid-switching (a) Mode 1

(b) Mode 2

(a) charging (IGBT: ON / FET: OFF)

(c) Mode 3

(b) charging (IGBT: OFF / FET: ON) (d) Mode 4 Fig. 3. Charging Operation in LLC Resonant Converter power transfer occurs during this mode. When SaH turns off and the secondary current starts to flow through the diode in SaL, ZVS occurs in SaL. The energy in resonant capacitor Cr is discharged. In mode 3, IGBT switch SaH turns off and IGBT switch SaL turns on as shown in Fig. 3(c). The primary current flows through the DC capacitor Cdc, the magnetizing inductance Lp, and the transistor in SaL. The secondary current flows through the leakage inductance Lr, the diode in SbH, the resonant capacitance Cr, and the magnetizing inductance Ls. The source power in primary side is transferred to the secondary side, and the resonant capacitor Cr is charged. In mode 4, both IGBT switches SaH and SaL turn off as shown in Fig. 3(d). The primary current flows through the diode in SaH, the DC capacitor Cdc, and the magnetizing inductance Lp. The secondary current flows through the leakage inductance Lr, the diode in switch SbH, the resonant capacitance Cr, and the magnetizing inductance Ls. No power transfer occurs during this mode. As the current

(c) discharging (IGBT: ON / FET: OFF)

(d) discharging (IGBT: OFF / FET: ON) Fig. 5. Hybrid-switching in 2-Quadrant Chopper

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through SaL turns off and the secondary current starts to flow through diode in SbH, ZVS occurs in SbH. The energy in resonant capacitor Cr is discharged.

2.3. Hybrid switching chopper Considering high efficiency and simple control, the LLC resonant converter operates with constant duty ratio and the 2-quadrant hybrid-switching chopper operates with variable duty ratio. Generally, IGBT has good performance in turn-on and conduction period, while FET has good performance in turn-off period. The IGBT tail current during the turn-off period can be removed by hybrid switching, which offers reduction of the switching loss. Fig. 4 shows the gating pulses for hybrid switching in the 2-quadrant chopper. The shunt-connected FET removes the IGBT tail current during turn-off period. Since the FET is only involved in turn-off period, the duration of gating pulse is about 2µs and the instant current rating is about 3 or 4 times normal rating. Also, the switching loss can be reduced about 2%. In case of charging, the IGBT switch for charging turns on while the FET for charging remains off-state in Fig. 4(a). The IGBT switch for charging turns off 2µs ahead the original turn-off instance. The FET turns on right before the turn-off instance of IGBT in Fig. 4(b). In case of discharging, the IGBT switch for discharging turns on, while the FET for discharging remains off-state in Fig. 4(c). The IGBT for discharging turns off 2µs ahead the original turn-off instance. The FET turns on right before the turnoff instance of IGBT in Fig. 4(d).

Fig. 6. Equivalent Circuit for Resonant Converter The IGBT collector-emitter capacitance Cce varies according to the magnitude of collector-emitter blocking voltage Vce. The relationship can be expressed by the following equation.

Cce (Vce ) = Co Vo / Vce

(1)

Where, Co is the value of Cce at the experimental voltage V o. Also, the diode junction capacitance Cj has identical characteristic. According to Eq. (1), the collector-emitter capacitance Cce increases sharply as the drain-source voltage Vce, that is line voltage decreases. Fig. 6 shows an equivalent circuit to determine the value of magnetizing inductance Lm for minimizing the loss. Where, the Cr and Cdc are much larger than the value of Cce and Cj . The switching loss of LLC resonant converter at n-th switching period can be obtained by the following equation.

2.4. High-frequency transformer design The design of high-frequency transformer is very important to reduce the switching loss and to implement the soft-switching. A section-type high-frequency transformer was chosen to utilize the leakage and magnetizing inductance. So, additional inductor connected in the secondary side can be removed, which is very critical to determine the size of DC-DC converter. In the LLC resonant converter, ZVS operation is closely related to the collector-emitter capacitance Cce of primaryside IGBT switch, the diode junction capacitance Cj of secondary-side IGBT switch, and magnetizing inductance Lm. Therefore, it is critical for improving the performance and efficiency of LLC resonant converter to determine the magnetizing inductance properly. Since the IGBT collector-emitter capacitance Cce becomes larger in the low line-voltage, the magnetizing inductance Lm should have small value for ZVS operation in whole range of input voltage. However, too small value of Lm brings about large conduction loss. So, in order to minimize the total loss, the value of Lm has to be determined by trading off the switching loss with the conduction loss.

Psr =

4 Cce (Vin [ n])vin [ n]2 K r Tsr

(2)

Where, kr is the total number of switching during a halfperiod of line voltage. If the starting voltage of ZVS is same as the input voltage Vin and its phase angle is θ , the number of switching from zero to θ is expressed by the following equation.

xn = int( kθ / π )

(3)

So, the switching loss during a half period of line voltage can be represented by the following equation. n = xn

Psr = ∑ n =1

8 Cce (Vin [ n])vin [n]2 kTs

(4)

If the switch turns off at the maximum value of primary-

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side magnetizing inductance and the switching period is defined by tfr, the turn-off loss can be defined by the following equation. n = kr

Poffr = ∑ n =1

Ts t fr Vin [n]2 kTLink 4 Lm

discharging, a non-linear model for was derived based on the charging and discharging characteristic curves. The system parameters of proposed power converter are described in Table 1 and the circuit parameters for DC-DC converter are described in Table 2. Fig. 7 shows the simulation results to verify the operation of proposed converter. Fig. 7(a) shows the phaseA voltage and current, and the DC link voltage during charging mode. The input voltage is in phase with the input current, which means that the power factor correction is properly operated. And the DC link voltage is maintained as a constant value of 700V. Fig. 7(b) shows the phase-A voltage and current, and the DC link voltage during discharging mode. The input voltage is 180 degree out phase with the input current, which means that the polarity of input current is negative during discharging. And the DC link voltage is maintained as a constant value of 700V. So, it is clear that the proposed converter can operates properly under accurate charging and discharging control. Fig. 8 shows the operation waveforms of LLC resonant converter. It is known that IGBT switch properly operates with a switching frequency of 50kHz. The primary and secondary resonant currents have sinusoidal waveforms without severe noise and ringing due to the high-frequency switching.

(5)

The conduction loss at the n-th switching period can be represented by the following equation. Pcr [n] = 2 I Lrp , rms [n]2 Rdsr / kr

(6)

Where, Rdsr is the conduction resistance of FET switch. The average conduction loss during a half-period of line voltage can be represented by the following equation. Pcr =

n = kr

∑ 2I n =1

in , rms

[n]2 Rdsr ] / kr

(7)

During the dead-time Tdead, the magnetizing current is used for the displacement current in resonant capacitor and the ZVS current of primary-side switch. Therefore, the magnetizing inductance Lm for ZVS operation under the given input voltage Vin can be expressed by the following equation. Lm =

Tdead 16 f s [Cce (Vin ) + (C j (Vin* / nT ) / nT 2 ]

(8)

*

Where, the resonant frequency fr is determined same as the switching frequency fs.

3. Simulation Analysis

(a) Charging

The performance of proposed power converter was verified by simulations with PSCAD/EMTDC. In order to analyze the battery operation during charging and Table 1. System Parameters of Proposed Charger Input voltage[L-L]

380V

Output voltage

450V

Output current

22.3A

Maximum output power

10kW

Buck boost converter switching frequency

25kHz

Resonant converter switching frequency

50kHz

(b) Discharging Fig. 7. Simulation Waveform of Grid-tied Inverter

Table 2. Circuit Parameters of Proposed Charger Transformer turn-ratio Secondary magnetizing inductance Ls Leakage inductance Lr Resonant capacitor Cr 2-quadrant chopper inductance L1 3-phase inverter inductance L

21 : 15 47.11uH 18.14uH 0.08uF 2.5mH 3mH

Fig. 8. Simulation Waveform of LLC Resonant Converter

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phase-A voltage and current, the phase angle reference signal, and the DC link voltage in charging mode. The input voltage is in phase with the input current, which means that the power factor correction is properly operated. And the DC link voltage is maintained as a constant value of 700V. It is clear that the PLL (phase-locked loop) operates accurately for converter control. Fig. 11(b) shows the phase-A voltage and current, and

Fig. 9. Operation Analysis of 2-Quadrant Chopper Fig. 9 shows the operation waveforms of 2-quadrant chopper. The DC link voltage shows transient phenomena at the instance of power flow reversal.

(a) Charging

4. Hardware Experiment A prototype of 10kW power converter for battery energy storage was built with commercially available components as shown in Fig. 10. The prototype was connected to the 3phase 380V line voltage and coupled with the lead-acid battery for charging and discharging experiments. Fig. 11 shows the experimental results to verify the operation of proposed converter. Fig. 11(a) shows the

(b) Discharging Fig. 11. Experimental Waveform of Grid-tied Inverter

(a) Switch voltage and resonant current in primary side

(b) Switch voltage and resonant current in secondary side Fig. 10. Hardware Prototype with 10kW Rating

Fig. 12. Experimental Waveform of LLC Resonant 1405


the DC link voltage in discharging mode. The input voltage is 180 degree out phase with the input current, which means that the polarity of input current is negative during discharging. And the DC link voltage is maintained as a constant value of 700V. So, it is clear that the proposed converter can operates properly under accurate charging and discharging control. Fig. 12 shows the operation waveforms of the LLC resonant converter, in which the voltages across the resonant capacitor Cr in the primary and secondary side are shown, and the resonant current in the primary and secondary side are also shown. It is known that IGBT switch properly operates with a switching frequency of 50 kHz. The resonant currents have sinusoidal waveforms without severe distortion and ringing due to the highfrequency switching. Fig. 13 shows the operation waveforms of hybridswitching in 2-quadrant chopper. Fig. 13(a) shows the voltage and current waveforms of IGBT and FET, which confirms the hybrid-switching behavior. The 2-quadrant chopper can operates with the switching frequency of 25 kHz without any problem. Using the hybrid switching scheme the switching loss can be reduced by about 2%. Fig. 14 shows the measured efficiency of proposed power converter with respect to the battery output power in the charging and discharging operation. Generally the efficiency of power converter rises as the output power goes up. In the rated power of 10kW, the efficiency is about 92.5% in discharging and 92% in discharging.

Fig. 14. Efficiency Analysis of Proposed DC-DC Power Converter

5. Discussion The AC-side harmonics of the proposed power converter is almost same as that of the existing power converter, but the battery-side ripple current is much lower because the DC-DC converter operates with high frequency switching. The size of 3-phase 60Hz 10kVA transformer is about 38cm x 17cm x 36cm that is equal to 22356cm3, whereas the size of 50kHz 5kVA high-frequency transformer is about 10cm x 10cm x 12cm that is equal to 1200cm3. Since two units are connected in shunt to handle 10kVA rating, total size is 2400 cm3. The size of each coupling reactor is about 12cm x 12cm x 16cm that is equal to 2304cm3. Total size for three units is 6912 cm3. Therefore, the net size could be reduced down to 40%. The efficiency of non-isolated hard-switching DC-DC converter is about 90%. Considering the efficiency of linefrequency transformer is about 97%, the overall efficiency of the existing hard-switching power converter is about 87%. This value is about 5% lower than that of the proposed DC-DC power converter. Also, the efficiency of proposed soft-switching DC-DC converter is about 1.5% higher than that of other soft-switching converters, because the first stage operates in fixed duty and the second stage operates in hybrid-switching scheme.

(a) Operation voltage and current

6. Conclusion In this paper a new grid-tied power converter for battery energy storage was proposed, which consists of a PWM inverter and a 2-stage DC-DC converter. The 2-stage converter is composed of an LLC resonant converter connected in cascade with a 2-quadrant hybrid-switching chopper. The LLC resonant converter operates in constant duty ratio, while the 2-quadrant hybrid-switching chopper operates in variable duty ratio for voltage regulation.

(b) IGBT tail current in turn-off state Fig. 13. Experimental Waveform of Hybrid-switching

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The operation of proposed system was verified through computer simulations. Based on computer simulations, a hardware prototype was built and tested to confirm the technical feasibility of proposed system. The proposed system could have relatively higher efficiency and smaller size than the existing system.

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Acknowledgements [11] This work (Grant No. 00045468) was supported by Business for Cooperative R&D between Industry, Academy, and Research Institute funded Korea Small and Medium Business Administration in 2012 and KEPCO(Korea Electric Power Corporation).

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Do-Hyun Kim He received his B.S. degree in Electrical Engineering from Myongji University. He is currently pursuing his Ph.D. degree at Myongji University. His current research interests include Bidirectional Intelligent Semiconductor Transformer for Smart Grid. Yoon-Seok Lee He received his B.S. degree in Electrical Engineering from Myongji University. He is currently pursuing his M.S. degree at Myongji University. His re-search interests include power elec-tronics applications for distributed generation and microgrids.

Byung-Moon Han He received his B.S. degree in Electrical Engineering from Seoul National University, Seoul, Korea, in 1976, and his M.S. and Ph.D. degrees from Arizona State University, USA, in 1988 and 1992, respectively. He was with the Westinghouse Electric Corporation as a Senior Research Engineer in the Science and Technology Center, Pittsburg, PA, USA. He is currently a Professor in the Department of Electrical Engineering, Myongji University, Seoul, Korea. His current research interests include power electronics applications for FACTS, custom power, distributed generation, and microgrid. Ju-Yong Kim He received his B.S, M.S and Ph.D. degrees in Electrical Engineering from Kyung-book University. He is currently Principal Researcher in KEPCO.

Woo-Gyu Chae He received his B.S. degrees in Electrical Engineering from Sung-kyun-kwan University and M.S. degrees in Electrical Engineering from Chung-book University. He is currently Senior Researcher in KEPCO.

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