High-Voltage, High-Performance Switch Using Series ... - IEEE Xplore

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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 25, NO. 9, SEPTEMBER 2010

High-Voltage, High-Performance Switch Using Series-Connected IGBTs Carmine Abbate, Giovanni Busatto, and Francesco Iannuzzo, Member, IEEE

Abstract—A high-performance, high-voltage switch obtained by a series connection of 3300 V insulated gate bipolar transistor (IGBT) modules is presented. The correct voltage sharing between the devices is ensured by the presence of a simple and reliable auxiliary circuit, which acts as a feedback on the gate terminal. A comparison in terms of power losses, output currents, and switching performances to a single 6500 V IGBT module is also presented. An extended experimental analysis demonstrates strong advantages in terms of switching losses (up to 67%) and current and voltage gradients. Use of trench-gate IGBTs, available in the 3300 V size but not in the 6500 V one, allows extra performance to the series connection circuit, as experimental results show. Such circuit is suitable to increase performances of high-power, high-voltage inverters used in railway traction, in high-voltage energy distribution lines, and in high-power solid-state klystron modulators. Index Terms—.

NOMENCLATURE Ca Cb Cin di/dt dv/dt EOFF EON Err Ic Ig Iload Lload N R Ra Rb Rg , OFF Rg , ON V∗ Vce Vg e Vg g , OFF Vg g , ON

Auxiliary capacitance [F]. Auxiliary capacitance [F]. IGBT input capacitance [F]. Current gradient [A/s]. Voltage gradient [V/s]. Turn-OFF Energy [J]. Turn-ON Energy [J]. Diode reverse-recovery energy [J]. Collector current [A]. Gate current [A]. Load current [A]. Load inductance [H]. Number of series-connected devices. Feedback resistance [Ω]. Auxiliary resistance [Ω]. Auxiliary resistance [Ω]. Gate off resistance [Ω]. Gate on resistance [Ω]. IGBT Miller voltage [V]. Collector–emitter voltage [V]. Gate–emitter voltage [V]. Negative IGBT driver voltage [V]. Positive IGBT driver voltage [V].

Manuscript received June 24, 2009; revised January 20, 2010; accepted April 11, 2010. Date of current version September 17, 2010. Recommended for publication by Associate Editor E. Santi. The authors are with the Department of Automation, Electromagnetism, Information Engineering, and Industrial Mathematics, University of Cassino, 43-03043 Cassino, Italy (e-mail: [email protected]; [email protected]; [email protected]). Digital Object Identifier 10.1109/TPEL.2010.2049272

∆Q ∆t ∆VOV τ ss τ ini τC b

Charge variation [C]. IGBT delay time [s]. IGBT over voltage [V]. Steady-state time constant [s]. First transient time constant [s]. Cb charge time constant [s]. I. INTRODUCTION

RESENTLY, the maximum insulated-gate-bipolar transistor (IGBT) rated voltage for commercial modules is 6500 V with a maximum continuous current of 600 A. Such devices are still in planar technology, and they cannot take benefits from the superior performances given by the trench-gate technology. So, series connection of IGBT becomes an interesting method to satisfy high-power converters designer requests [1]–[3]. In fact, thanks to the series connection techniques, higher operating voltage can be reached having, at the same time, the high performances of devices in terms of safe operating area (SOA) and switching losses. Finally, the series connection could result in a real improvement of the total weight, volume, and cost of the whole converter, thanks to the possibility of both reducing energy losses and increasing operating frequency. Moreover, in the series connection applications, lower voltage, higher performances, latest technology devices can be used in order to obtain the maximum switching performances. Nevertheless, the series connection introduces several problems related to the dynamic voltage sharing across the devices. In effects, timing shifts of the driver circuits as well as physical differences in the structure of the devices can induce uneven distribution of the overall voltage across the series-connected IGBTs that could result in failure. Large efforts have been dedicated in literature to electronics topologies, which allow the safe series connection of power devices [1]–[5], but only in recent years, the series connection techniques have been extended to the high power IGBT modules used in railway applications [2]. A simple and reliable auxiliary circuit has been introduced in [1], [2] that ensures the system reliability requirements that, instead, are conditioned and endangered by the complex analogue-digital-balancing circuit usually proposed in literature [4]–[8]. The circuit topology used in this paper was proposed in the past [1] and now designed and optimized for high-power devices. In fact, in high-power applications, we must consider the problems related to the very high current value and to the delay time introduced by the power device [2]. The application of such series connection technique to 1700 V-800 A IGBT modules proposed in [2], compared to the single 3300 V IGBT module, has shown very good

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ABBATE et al.: HIGH-VOLTAGE, HIGH-PERFORMANCE SWITCH USING SERIES-CONNECTED IGBTs

Fig. 1.

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Electric schematic of proposed circuit. C a , C b , R a , R b , D, and R are the auxiliary balancing circuits. (Inset) Detail of the driver front end.

advantages in terms of power losses, switching frequency, and system robustness [9]. The aim of this paper is to obtain a high-voltage low-loss switch, using series-connected 3300 V-1200 A IGBT modules, in planar and trench technologies. Moreover, a comparison in terms of power losses, switching performances, and inverteroutput current with a single 6500 V IGBT module is presented. The analysis is also performed using latest generation 3300 V trench-gate IGBT modules [10], [11]. An extensive experimental analysis, executed on a phase-leg assembly, demonstrates strong advantages of the series-connected IGBTs in terms of power dissipation, devices’ overvoltages, switching frequency, and maximum load current.

II. EXPERIMENTAL SETUP AND OPERATION PRINCIPLE The electrical schematic of the phase-leg assembly, using the proposed voltage-balancing circuit is reported in Fig. 1. As shown in the figure, an auxiliary circuit, composed of diodes and passive resistive and capacitive components, is inserted between the collector and the emitter of each IGBT module. For the sake of clarity, the schematic of one of the IGBT drivers is also reported in the inset of Fig. 1. The circuit was proposed in [1] for low power devices, and in this paper extended for high power IGBT modules. The operation of the auxiliary circuit is reported in [1]. Here, we want just to recall very briefly its basic operation and present some considerations regarding the static and dynamic sharing of the

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voltage among the series-connected devices not fully described in [1]. The circuit basically operates on the gate lead: when the collector–emitter voltage of a given IGBT exceeds the reference voltage across its precharged Ca capacitance, an additional current is forced on the gate terminal through the components D and R, thus introducing a negative feedback. In this way, during both turn-ON and turn-OFF transients, the circuit drives the devices that would tend to be in the OFF state, i.e., the slowest and the fastest ones at the turn ON and OFF, respectively, and keeps them in their active region for the whole time extent of the voltage unbalance. As it was explained in [2], despite additional switching losses with respect to the case of a single device, the use of lower voltage more performing series-connected devices guarantees an overall loss reduction. The way to how the reference voltage on Ca tends to Vcc /N at the steady-state condition was only partly discussed in [1] with reference to a step-down configuration. First, the steady-state uniform distribution of the voltage is the result of a transient phenomenon, which takes place once the voltage is applied for the first time on the series-connected devices. This phenomenon is fast and involves the capacitor Ca in series with the related R, Rg , OFF , and the negative driver power supply Vg g , OFF , as discussed in Appendix A. Second, due to the passive nature of the components, the uniform distribution of the voltage among each section can easily be guaranteed during the whole transient evolution, thanks to a proper choice of their values, as shown in Appendix A. The intrinsic modularity of the circuit allows to extend the topology to several IGBT devices in order to obtain a very highvoltage switch [1]. Thanks to the presence of passive resistive-capacitive elements, the circuit used in this paper can result much more reliable [9] with respect to other proposed strategies that, in general, make use of complex digital logic [4]–[7]. For this reason, the circuit can be used in power applications, as in railway ones, where the reliability is a strong limitation. Moreover, the circuit can be implemented on existing inverters with modest modifications and, consequently, low adjustment costs. Furthermore, thanks to the presence of a gate-feedback circuit, the auxiliary components have smaller dimensions and costs, if compared with traditional residual-current device snubbers. Finally, as it has been demonstrated in [9], the proposed circuit exhibits very good performances for diode operations too, allowing to obtain a good equivalent high-voltage diode. III. TURN-OFF TESTS Two device part numbers have been successfully used for tests, rated at 3300 V, 1200 A. The first one is a second generation, punch through device, whereas the second one is a third generation, trench-gate IGBT. The obtained results have been compared with the ones of a single 6500 V, 600 A second generation, planar IGBT module. In fact, at this time, the trench-gate technology is not available for 6500 V modules; besides in 6500 V size, the maximum nominal current is 600 A. The first step has been the design of the auxiliary circuit. Simple design

rules have been followed for the choice of the components, in order to optimize the devices’ performance in terms of additional switching losses and collector overvoltage. In fact, as reported in [3], a tradeoff between the maximum collector-voltage value and the additional losses must be considered during the design. The main equations used for the calculation of the auxiliary components are reported in Appendix A. The circuit in Fig. 1 represents an inverter leg and is subdivided in four identical sections, two for the low-side switch and two for the high-side one. In particular, in our tests, sections 1 and 2 operate as two series-connected freewheeling diodes. As stated before, in fact, the auxiliary circuit can also be used for series-connected diodes, where the voltage sharing among the diodes is assured by the conductivity modulation of the IGBT device connected to the faster diode [9]. The first experimental characterization has been executed on planar 3300 V (second generation) commercial IGBT modules. In order to guarantee the best performances of the modules, we used the gate resistances suggested by the manufacturer, namely Rg , ON = Rg , OFF = 1.2 Ω. Moreover, as suggested by the manufacturer, we added an external capacitor between the gate and the emitter leads in order to optimize the device commutation. The total input capacitance of the device was Cin = 370 nF. The values of the auxiliary circuit components, calculated with the procedure reported in [1] and in Appendix A, were: R = 30 Ω, Ca = 720 nF, Cb = 1 nF, Ra = 100 kΩ, and Rb = 9 kΩ. The Ca and Cb components are high voltage (Vcc /2 = 2000 V), low capacitance, polypropylene-film capacitors. The size of these components (4 × 2 cm) is very small if compared with the corresponding snubber capacitors. The diode D is a low current, fast, 3000 V rate diode obtained with the series connection of three diodes rated at 1000 V and 4 A. In order to simulate a bad synchronization between the devices, a delay between the two gate commands of the lower switch has been artificially set. From specifications of such devices, the maximum difference in delay time between devices is lower than 1.2 µs. Fig. 2(a) shows the turn-OFF waveforms in the case of a delay time of 700 ns. In particular, the gate-OFF command of the higher IGBT (section 3) has been delayed with respect to the lower one (section 4). The static supply is set to Vcc = 4000 V, and the Iload is 500 A. The other parameters are: gate-driver resistance Rg , ON = Rg , OFF = 1.2 Ω and load inductance Lload = 100 µH. Referring to Fig. 2(a), the collector current (Ic ), the lower IGBT voltage (QL Vce ), the higher IGBT voltage (QH Vce ), and the total voltage across the section 3 and 4 (total Vce ) have been reported. It is easy to observe that, even with such a large delay, the balancing circuit is able to ensure a good sharing of the voltage between the two modules: the maximum measured voltage is lower than 2800 V. The auxiliary circuit ensures the voltage sharing by keeping the faster IGBT device in its conductivity modulation region during its turn-OFF. The operating principle is shown in Fig. 2(b), where the gate voltage of the IGBT of section 4 is reported. It can be observed that due to the presence of Ca , D, and R components, an additional current is forced into the gate terminal when the collector voltage on the device exceeds the reference voltage. This current increases the gate voltage and


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Fig. 4. Turn-OFF waveforms of two series-connected 3300 V, trench-gate IGBT modules: collector current and voltages.

Fig. 2. Turn-OFF waveforms of two series-connected 3300 V, secondgeneration IGBT modules: (a) collector current and voltages and (b) corresponding low-side gate voltage.

Fig. 3. Single 6500 V IGBT module: collector current and voltage turn-OFF waveforms.

maintains the IGBT in its active region during all the delay time, making the device to be subjected to the additional power losses that increase its operating temperature. The commutation has been compared with one of a single 6500 V IGBT module in standard driving conditions, whose waveforms are reported in Fig. 3. In this test, a much lowervoltage slope can be recognized. In particular, the voltage slope is 8.1 kV/µs for the series connection and 2.2 kV/µs for the single 6500 V device, and consequently, a much larger dissipation is observed. The turn-OFF energy, calculated by means of the procedure reported in Appendix B, was EOFF = 2400 mJ for the 6500 V IGBT, and EOFF = 800 mJ for the two series-connected

second-generation IGBT modules with a perfect synchronization. For the real performances comparison, we have considered the case of perfect device synchronization: the turn-OFF energy for the series connection is reduced by a factor of 3. The experimental characterization also demonstrates that the series-connected IGBTs exhibit a lower collector-voltage peak (see Figs. 2(a) and 3, respectively). Such desirable achievement is due to the auxiliary circuit that also reduces the collector overvoltage during turn-OFF. In order to compare the performances of series-connected IGBT modules during turn-OFF operations, the experimental characterization has been also executed on last third-generation 3300 V trench-gate IGBT modules. The input capacitance of the devices, considering also the external one, is: Cin = 310 nF, whereas the ON and OFF-gate resistances, suggested by the manufacturer are: Rg , ON = 1.3 Ω, Rg , OFF = 5.6 Ω. The values of the auxiliary circuit components, calculated with the procedure reported in Appendix A, are: R = 100 Ω, Ca = 220 nF, Cb = 1 nF, Ra = 250 kΩ, Rb = 25 kΩ. The values of the Ca capacitors are smaller than those obtained for the previous IGBT modules: Ca and R values are directly related to the device input capacitance and OFF resistance (see Appendix A). Fig. 4 shows the turn-OFF collector current and voltage waveforms for series-connected trench IGBT, measured with a delay time of 700 ns. The static-supply voltage is set to 4000 V, and the load current to 500 A. The other parameters for the tests are: gate-driver resistances Rg , ON = 1.2 Ω, Rg , OFF = 5.6 Ω, and load inductance Lload = 100 µH. Again, a very good voltage sharing is ensured by the auxiliary circuit. With reference to Fig. 4, the collector current (Ic ), the voltage across the lower IGBT device (QL Vce ) and the higher IGBT (QH Vce ), and the total voltage across sections 3 and 4 (total Vce ) have been reported. Moreover, in Fig. 4, lower voltage peaks with respect to second-generation IGBT modules can be observed, even if the collector-current gradient is lower than the previous case. The performances of the series-connected 3300 V-1200 A IGBT modules during the turn-OFF with respect to the case of a single IGBT are confirmed by the calculation of the energy losses. In fact, Fig. 5 reports the turn-OFF energy losses, measured at 25 ◦ C, as a function of the delay time. The energy loss has been calculated at Vcc = 4000 V and Ic = 500 A. As reported

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Fig. 5. Turn-OFF energy losses versus delay time at Ic = 500 A for a single 6500 V IGBT device, two series-connected second-generation devices and two trench-gate devices.

Fig. 6. Turn-ON waveforms of two series-connected 3300 V IGBT module: collector current and voltages.

in [2], the turn-OFF energy is approximately linearly related to the delay time. In fact, as detailed in the earlier paragraph, the faster IGBT module is maintained in its active region during the whole delay time. Considering the linearity of the OFF energy with the delay time, we can estimate that the series connection of planar IGBT modules is always competitive with respect to the single device for all the range of delay time (0 ÷ 1.2 µs); instead trench IGBTs are competitive for delay times below 600 ns. In the ideal case of perfect synchronization between the devices (delay time = 0), the lower OFF-energy is obtained with the two 3300 V planar IGBT modules. In fact, the turn-OFF energy losses for planar IGBTs are reduced to about 67% (from 2.4 to 0.8 J), while for trench-gate devices, they are reduced to about 37% (from 2.4 to 1.5 J). Of course, for the sake of comparison, we must also consider the turn-ON losses and the static losses that will be calculated in the following paragraph. IV. TURN-ON TESTS Switching performances have been also investigated and compared at the turn-ON transient. The test conditions and the parameter values are the same specified for the turn-OFF tests (see Section III). Fig. 6 shows the collector current and voltage waveforms at the turn-ON, measured in the case of a delay of 600 ns. The load current is fixed to 500 A, while the dc voltage is 4000 V. In particular, the low-side IGBT (section 4 in Fig. 6) has been

Fig. 7. Turn-ON waveforms of a single 6500 V IGBT module: collector current and voltages.

delayed with respect to the high side one. In this way, the lowside IGBT experiences a collector overvoltage. Although very high delay time has been imposed, the balancing circuit is able to maintain the maximum collector voltage lower than 2700 V (see the QL Vce curve in Fig. 6). The voltage reduction is ensured by the conductivity modulation of the low-side IGBT module during the whole delay time. The waveforms reported in the same figure show also very high current and voltage gradients. The current peak on the collector current is related to the reverse recovery of the freewheeling series diodes. A small oscillation phenomenon, due to the low value of the feedback resistor, on the collector current and voltage can be recognized. The same test has been repeated on the single 6500 V device. The corresponding turn-ON collector waveforms are reported in Fig. 7. The waveforms of 6500 V IGBT show lower values of voltage and current gradients that significantly increase the dissipated energy, even if a lower peak on collector current is produced. The turn-ON tests have been executed also on the trenchgate IGBT modules. For example, Fig. 8(a) reports the collector waveforms measured at Ic = 550 A, Vcc = 4000 V, and a delay time of 700 ns. The other parameters of the test are: Rg , ON = 1.2 Ω, Rg , OFF = 5.6 Ω, and load inductance Lload = 100 µH. Thanks to the presence of the auxiliary circuit, the maximum collector voltage is limited to 2900 V. The feedback voltage on the gate side is depicted in Fig. 8(b) where the gate-voltage evolution is reported. We can see that, just in correspondence of the voltage collector peak (at about 1.4 µs), the gate voltage is forced to increase to 15 V. In this way, the IGBT voltage is reduced very quickly. The collector-current peak value related to the freewheeling diode is the same of 3300 V planar IGBT series connected. Also, in this case, the collector gradients are very high if compared with 6500 V-600 A IGBT modules. No oscillating phenomena have been observed. Fig. 9 reports the turn-ON energy losses of single 6500 V, planar and trench IGBT modules as a function of devices’ delay time for a load current of 500 A, measured at 25 ◦ C. As shown in the figure, the total losses are practically independent of the delay time. This behavior can be associated to the fact that the current through the series-connected modules is conditioned by


Fig. 8. Turn-ON waveforms of two series-connected 3300 V trench-gate IGBT modules: (a) collector current and voltages and (b) corresponding low-side gate voltage.

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Fig. 10. Turn-ON waveforms of two series-connected 3300 V trench-gate IGBT module at Ic = 1200 A and V c e = 4000 V: (a) collector current and voltages and (b) corresponding switching loci on RBSOA.

V. RATING OF CONVERTERS USING SERIES-CONNECTED IGBT MODULES

Fig. 9. Turn-ON energy losses versus delay time at Ic = 500 A: comparison between single 6500 V IGBT, 3300 V second-generation IGBT, and 3300 V trench devices.

the slowest one, which keeps low current flow through it in the first part of the transient until the auxiliary circuit forces it to turn on. Consequently, the contribution of this first part to the energy losses is negligible and the losses are related only to the current rise induced by the slowest device and to the diodes reverse recovery. Thanks to the analysis of Fig. 9, we can conclude that the turn-ON losses for IGBT series connected are considerably lower (∼60%) than 6500 V single device and quite similar between the two 3300 V trench-gate IGBTs and the two planar ones.

In this section, we discuss some considerations about the usage of series connection of two IGBT modules in substitution of a single 6500 V IGBT module. In typical heavy traction application, the nominal load current is not lower than 1200 A. This means that the basic component in every inverter leg requires two 6500 V-600 A IGBT modules parallel connected or two 3300 V-1200 A series connected. The consequence is that it is convenient to use series connection, as it does not increase the number of devices required in the inverter and at the same time presents the good advantages shown in the previous paragraphs. In particular, the main recognized advantages are in switching losses reduction and in dynamic gradients improvements. In order to confirm this convenience, the experimental analysis has been extended at higher load currents. Fig. 10(a) depicts the voltage and current waveforms at Ic = 1200 A and Vcc = 4000 V for the trench IGBT modules, series connected. The delay time has been set to 600 ns. Also, in this case, the auxiliary circuit acts in order to reduce the collector overvoltage. In particular, the maximum-collector voltage is about 2400 V, similar to the tests at 500 A. The corresponding switching loci of the low-side and the high-side IGBT modules are reported in Fig. 10(b), together with the reverse-bias SOA (RBSOA). We can see that the voltage and current are well inside the safe boundaries.

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In particular, the total inverter losses include the IGBT and diode static losses, the IGBT and diode switching losses, and the losses related to the delay time. In order to calculate the IGBT switching losses, we have considered (1) Psw I G B T = fsw (EON + EOFF )

Fig. 11. Turn-ON waveforms of two series-connected 3300 V trench-gate IGBT modules at Ic = 1200 A: collector current and voltages.

where fsw is the working switching frequency, EON and EOFF are the turn-ON and the turn-OFF energies, measured at turnOFF reference voltage (V nom OFF ) and current (Inom OFF ). Moreover, Iav I G B T and Iav D i o d e represent the average IGBT and diode currents. Regarding the diode switching losses, a similar (2) can be used Psw D i o d e = fsw Err

Fig. 12. Comparison between static voltage drops for series-connected IGBTs and single 6500 V device: V g e = 15 V, T s = 125 ◦ C.

A similar analysis has been repeated during the turn-ON operations. The respective collector waveforms are reported in Fig. 11. The other parameters for the tests are: collector current Ic = 1200 A, Vcc = 4000 V, load inductance Lload = 100 µH, and delay time ∆t = 700 ns. The maximum-collector voltage, measured on the low-side IGBT (section 4 in Fig. 11) is about 2750 V. Although the very high delay time, the auxiliary circuit ensures a very good voltage sharing. Another advantage in using series-connected trench IGBTs comes from their static ON-state voltage drop. Fig. 12 shows the static characteristics, measured at 125 ◦ C base plate temperature, for a single 6500 V IGBT module (line with “o”), the series connection of two 3300 V second-generation IGBT (line with “x”), and the series connection of two trench-gate IGBT modules (solid line). We can see that, for a given collector current, the ON-voltage drop for two trench IGBT modules is even lower than that of a single 6500 V IGBT, as expected from the trenchgate technology that allows to obtain fast commutations and at the same time, low ON-voltage drop. On the other side, the ONstate voltage of series-connected 3300 V planar IGBT is larger than one of the single 6500 V as one could expect. On the basis of the earlier results, a comparison between the performances of two series 3300 V, planar and trench technologies, and two parallel 6500 V IGBTs in a real three-phase heavy-traction inverter is now presented. For this purpose, we numerically evaluated the maximum value of the rms output current that can be achieved by using those modules.

Vdc Iav I G B T + Iav D i o d e (1) Vnom OFF Inom OFF

Vdc Vnom OFF

Iav I G B T + Iav D i o d e Inom OFF

(2)

where Err is the diode reverse-recovery energy, measured at a reference voltage and current. The basic approximation in (1) and (2) is that the IGBT and diode switching losses are linearly related to the working voltage and current values [12]. In this way, a linear voltage and current losses scaling, with respect to a reference value, can be executed. Moreover, the EON , EOFF , and Err energies, used as reference values, can be measured at a junction temperature near the operating one. The additional losses introduced by the proposed auxiliary circuit, and related to the delay time, must be considered. We can remember that, at the turn-OFF, these losses are practically linearly dependent on the delay time, the voltage, and the current values [2], as has been confirmed by earlier experimental tests. Moreover, the additional turn-ON losses are independent of the delay time. Starting from the previous considerations and approximations, we have computed the devices’ temperature by using a standard thermal model of the inverter fed by the measured power losses in presence of the auxiliary circuit. The final step is to numerically calculate the maximum output current admissible for a specified maximum operating junction temperature. We used two different maximum junction temperatures: 135 ◦ C for trench and 110 ◦ C for planar IGBTs. For all cases, we held a 15 ◦ C margin with respect to the maximum junction temperatures set by manufacturers to 150 ◦ C and 125 ◦ C for trench and planar IGBTs, respectively. Fig. 13 shows the results of the aforementioned analysis. It reports the theoretical admissible output rms current for a heavy traction inverter where different combinations of modules are used: 1) 2 × 6500 V parallel-connected IGBTs (solid line); 2) 2 × 3300 V series-connected second-generation IGBTs (line with “x”); and 3) 2 × 3300 V series-connected trench IGBTs (line with “o”). As in the other cases, the results are reported as a function of the delay time. In our reference inverter, the dc voltage has been set to 4000 V and the switching frequency to 750 Hz. The curves demonstrate that using trench IGBTs allows to achieve output currents larger by a factor of 2 than using 6500 V modules. The advantage for the series connection remains high up to a delay time of 1 µs.


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Iload and the device operates in the Miller phase. This additional current flowing through Rg , OFF is Ig =

V ∗ − Vg g ,oﬀ . Rg ,oﬀ

The feedback resistance R can be calculated as ∆VOV − V ∗ R= . Ig

Fig. 13. Comparison among traditional and trench IGBT series connected in inverter applications.

(A1)

(A2)

In our project, the maximum over-voltage on the 3300 V IGBTs has been set to 500 V. The value of the capacitance Ca must be defined in order to keep practically constant the voltage across it during the duration of the whole dynamic unbalancing condition. In particular, the charge that Ca must supply to the gate circuit is

VI. CONCLUSION

∆QOFF = Ig ∆t

In this paper, we have demonstrated that the series connection of the lower voltage IGBTs shows advantages also for traditional technologies (second-generation IGBT in our example). In conclusion, we can extend the series connection techniques on all the IGBT technologies, in order to obtain the losses, weight, and cost reduction of power converters [13]–[16]. The voltage sharing across the devices has been obtained using a simple and reliable circuit that has been proposed in the past and now optimized for high-power applications. Its performances have been compared with those of a single 6500 V600 A IGBT device in standard-driving conditions, in terms of power dissipation and switching performances. An experimental analysis has been also performed on trench-gate IGBT modules. The experimental tests show strong advantages both at turnOFF and turn-ON. In particular, the reduction in switching losses is up to 65% by using standard IGBT technology, but they can be further increased using latest technology trench-gate IGBT modules. The advantages of series-connection techniques have been also demonstrated in a typical inverter for heavy traction for which maximizing the output rms current is the main goal. The output current can be increased up to a factor of 2 by using series-connected trench-gate IGBT modules. The series connection technique can, in principle, be extended to several IGBT devices in order to obtain very high values of blocking voltage.

where ∆t is the duration of the dynamic unbalancing phase, which corresponds in our assumption to the delay between the series-connected device. The capacitance can be calculated as

APPENDIX A DIMENSIONING THE AUXILIARY CIRCUIT COMPONENTS The value of the components used in the auxiliary circuit can be calculated with some theoretical considerations based on the steady-state operation of the circuit. The first parameter to be defined is the resistor R whose role is to keep the device in its active region when an over-voltage, ∆VOV = (Vce − Vcc )/N , appears across it [1]. For doing that, while the gate driver tends to keep the gate voltage at Vg g , OFF , the auxiliary circuit is required to supply an additional current through Rg , OFF that feeds at the gate lead a voltage equal to V ∗ = G−1 (Iload ), with Ic = G(Vg ) the transconductance of the device. In other words, V ∗ is the gate voltage for which Ic =

Ca =

∆QOFF ∆VC a

(A3)

(A4)

where ∆VC a is the accepted variation of the voltage across Ca . In our application, we fixed ∆VC a to be 4% of the nominal Vcc /2: ∆VC a = 50 V. The next parameter to be defined is Rb whose role is to guarantee that ∆VC a is reduced to zero during the subsequent ON-state phase of the series-connected IGBTs, in such a way to maintain the voltage across Ca stably at Vcc /N in the steady-state conditions. Let us consider that during the ON-phase the voltage across Ca evolves according to an exponential law with the time constant τss = Ca /Cb Ra /Rb that reduces to τss ∼ = Ca Rb due to the fact that Ra Rb and Cb Ca [1]. The minimum duration TON,m in of the ON phase must be significantly larger than τss . If we assume a ratio 5 between these two quantities, we can obtain Rb =

τss TON,m in = . Ca 5Ca

(A5)

It is worth outlining that, thanks to the choices performed earlier, the computed value of Rb is naturally larger than R. In fact, if we combine (A2), (A3), and (A4), and neglect V ∗ with respect to ∆VOV , we can obtain R∼ =

∆VOV ∆t Ca ∆VC a

(A6)

and consequently, combining (A5) and (A6), we can conclude Rb TON,m in ∝ . (A7) R 5∆t Typically TON,m in ∆t thus indicating that Rb R, moreover, R is of the same order of Rg , OFF so that we can conclude that Rb R + Rg , OFF as indicated in Section II. The other two parameters to be defined are Ra and Cb , which must be much larger than Rb and much lower than Ca , respectively [1]. A reasonable choice for these components is Ca ≈ 100 Cb

(A8)

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and Ra ≈ 10. Rb

(A9)

Regarding the dynamic operation of the circuit, we have to outline that it is dominated by the following key time constants. 1) τss = Ca /Cb Ra /Rb which reduces to τss ∼ = Ca Rb and defines the steady-state operation of the circuit during the ON phase of the series switch. 2) τini which regulates the first transient when the voltage on the capacitor Ca is charged to ≈Vcc /N . This transient involves the circuit made by Ca /Ra in series with Cb /Rb /(R + Rg , OFF + Vg g , OFF ). Because the good operation of the auxiliary circuit requires that Ra Rb R + Rg , OFF and Cb Ca , the currents through Ra , Cb , and Rb are negligible so that the N capacitors Ca are charged through the corresponding R, Rg , OFF , and Vg g , OFF and the time constant τini reduces to τini ∼ = Ca (R + Rg , OFF ). 3) τC b which defines the transient when the capacitor Cb are charged to the value ≈−Vg g , OFF at the beginning of the initial charge of the capacitor Ca . This transient starts when the Vcc voltage is applied for the first time to the seriesconnected switches and the voltage across the capacitor Ca is practically zero. The total Vcc is then applied on the series of the N equal circuits involving the driver and constituted by Cb /(Rb in parallel with R, Rg , OFF , and Vg g , OFF ). The time constant reduces τC b ∼ = Cb (R + Rg , OFF ) considering that Rb R + Rg , OFF . The resistive nature of the circuit in parallel to Cb guarantees the dynamic uniform distribution of the voltage among the various sections. During this transient, the voltage across each capacitors Cb starts from Vcc /N and exponentially becomes ≈−Vg g , OFF at the end of the transient. If we consider the assumption regarding the values of the passive components previously used, we can say that τC b τini τss , and, consequently, the voltage across the auxiliary circuit components reaches their steady state value quite rapidly. Moreover, the passive nature of these components helps in keeping always below Vcc /N the voltage across each section of the series-connected switches even during the start up of the converter and permits safe operations in any conditions. APPENDIX B ENERGY CALCULATION The values of the energies, reported in this paper, were calculated with a numerical procedure implemented directly by the waveforms acquisition software. In fact, the Tektronix oscilloscope TDS2024 used to acquire the waveforms, has been coupled by a GPIB acquisition system to a personal computer, which allowed us to calculate the dissipated energies during the commutation. The collector currents have been measured by a Pearson current sensor model 1025 and the collector voltages by a TEK P5100 high voltage probe. In order to reduce the measurement errors in power losses calculation, due to the time shift between acquired waveforms, an accurate probes compensation proce-

Fig. 14. Time evolution of dissipated energy during a typical turn-OFF on inductive load of an IGBT.

dure was executed before the experiment. The measurement of the cables delay has been executed by measuring, with the experimental setup, current and voltage waveforms during several commutations of a 3300 V IGBT on a noninductive resistive load. In fact, the voltage and current waveforms for these commutations should be complementary to each other and if we invert one of them it differs from the other only because of the probes delays. For reducing the measurement errors, we adjusted the length of the cable, which connects the current sensor to the oscilloscope until we reduced to less than 5 ns the value of the measured delay between current and voltage. Once the cable length was fixed, we verified that the mean value of this delay measured on ten commutations was below the 5 ns limit. The switching energy losses can be calculated as the time integral of the instantaneous power obtained by the product of the measured collector voltage and current time evolution, as indicated in the following [7]:

t2

Vce (t) Ic (t)dt.

E=

(B1)

t1

The typical time evolution of the instantaneous energy dissipated during an IGBT inductive turn OFF, that is the integral (B1), is reported in Fig. 14. The first linear smooth increase until t1 is related to the conduction phase of the IGBT where Vce = Vce ,sat and Ic = Iload both being practically constant. At the instant t1 the turn-OFF starts and the voltage rise causes the rapid increase of the curve slope. After the voltage rise is accomplished, the slope of the energy time evolution slowly reduces because of the current tailing. We can consider that the turn-OFF is completed at the instant t2 when the slope becomes zero and the curve becomes flat. We have used the time evolutions of the lost energy, like the one shown in Fig. 14 associated to each commutation, to measure the turn-OFF energy losses as the variation of the energy between the values that it assumes at the instants t1 and t2. In such a way, we accurately define the time integration limit, t1 and t2, and we accurately account for the total energy lost during the whole collector current tail. We used a similar procedure also for measuring the energy lost during the device turn-ON but we do not report it here for brevity.


REFERENCES [1] J. W. Baek, D. W. Yoo, and H. G. Kim, “High voltage switch using seriesconnected IGBTs with simple auxiliary circuit,” in Proc. Ind. Appl. Conf., Oct. 8–12, 2000, vol. 4, pp. 2237–2242. [2] C. Abbate, G. Busatto, L. Fratelli, F. Iannuzzo, B. Cascone, and G. Giannini, “Series connection of high power IGBT modules for traction applications,” presented at the Conf. EPE, Dresden, Germany, Sep. 11–14, 2005. [3] K. Sasagawa, Y. Abe, and K. Matsuse, “Voltage-balancing method for IGBTs connected in series,” IEEE Trans. Ind. Appl., vol. 40, no. 4, pp. 1025–1030, Jul./Aug. 2004. [4] C. Gerster, “Fast high-power/high-voltage switch using series-connected IGBTs with active gate-controlled voltage-balancing,” in Proc. 9th Annu. Conf. APEC, Feb. 13–17, 1994, vol. 1, pp. 469–472. [5] P. R. Palmer and N. Githiari, “The series connection of IGBTs with active voltage sharing,” IEEE Trans. Power Electron., vol. 12, no. 4, pp. 637– 644, Jul. 1997. [6] R. Withanage, W. Crookes, and N. Shammas, “Novel voltage balancing technique for series connection of IGBTs,” presented at the Conf. EPE, Aalborg, Denmark, Sep. 2–5, 2007. [7] D. V. M. M. Krishna and V. Agarwal, “Active gate control of series connected IGBTs using positive current feedback technique,” IEEE Trans. Circuits Syst. II, Exp. Briefs, vol. 52, no. 6, pp. 261–265, May 2005. [8] G. Belverde, A. Galluzzo, M. Melito, S. Musumeci, and A. Raciti, “Snubberless voltage sharing of series-connected insulated-gate devices by a novel gate control strategy,” IEEE Trans. Power Electron., vol. 16, no. 1, pp. 132–141, Jan. 2001. [9] C. Abbate, G. Busatto, L. Fratelli, F. Iannuzzo, B. Cascone, and R. Manzo, “The robustness of series-connected high power IGBT modules,” Microelectron. Reliability, vol. 47, pp. 1746–1750, 2007. [10] A. Pfaffenlehner, J. Biermann, C. Schaeffer, and H. Schulze, “New 3300 V chip generation with a trench IGBT and an optimized field stop concept with a smooth switching behavior,” in Proc. ISPSD, May 24–27, 2004, pp. 107–110. [11] M. Bakran, H. G. Eckel, M. Helsper, and A. Nagel, “Next generation of IGBT-modules applied to high power traction,” presented at the EPE Conf., Aalborg, Denmark, Sep. 2–5, 2007. [12] N. Mohan, T. M. Undeland, and W. P. Robbins, Power Electronics: Converters, Applications, and Design, 3rd ed. New York: Wiley, 2002. [13] G. Busatto, B. Cascone, L. Fratelli, and A. Luciano, “Series connection of IGBTs in hard-switching applications,” in Proc. IEEE Ind. Appl. Conf., 3rd IAS Annu. Meet., Oct. 12–15, 1998, vol. 2, pp. 825–830. [14] Y. Abe, K. Maruyama, Y. Matsumoto, K. Sasagawa, and K. Matsuse, “Performance of IGBTs series connection technologies for auxiliary power supply system,” in Proc. Power Convers. Conf., Nagoya, Japan, Apr. 2–5, 2007, pp. 1382–1387. [15] Y. Abe, K. Matsubara, T. Mochida, and K. Matsuse, “A novel method for loss reduction in high-voltage inverters,” in Proc. IAS, Oct. 2–6, 2005, vol. 3, pp. 1849–1854. [16] T. Kjellqvist, S. Ostlund, and S. Norrga, “Active snubber circuit for source commutated converters utilizing the IGBT in the linear region,” IEEE Trans. Power Electron., vol. 23, no. 5, pp. 2595–2601, Sep. 2008.

Carmine Abbate was born in Italy, in 1976. He received the degree in telecommunication engineering and the Ph.D. degree in electrical and information engineering from the University of Cassino, Cassino, Italy, in 2001 and 2006, respectively, where he was involved in a study on the electromagnetic interference generated of power insulated gate bipolar transistor modules in power converters. He is currently in the Department of Automation, Electromagnetism, Information Engineering, and Industrial Mathematics, University of Cassino. He is the author or coauthor of more than 30 publications on international conferences and journals. His research interests include in the area of power devices reliability, power devices analysis and characterization, innovative converter, and driving topologies and in electromagnetic interference generated of power converters. Dr. Abbate is a Reviewer of the IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS.

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Giovanni Busatto received the Laurea degree in electronic engineering from the University of Naples, Naples, Italy, in 1983. From 1984 to 1992, he was a Researcher at the Institute of the Italian National Council of Research. During 1987 and 1988, he was a Visiting Researcher at Thomas J. Watson Research Center, Yorktown Heights, NY. In 1992, he was a Visiting Researcher at the National Institute of Standard and Technology, Gaithersburg, MD. From 1992 to 1998, he was an Associate Professor of power electronics at the University of Naples “Federico II”. In 1998, he joined the University of Cassino, Cassino, Italy, as an Associate Professor of electronics, where he became a Full Professor of power electronics and telecommunication electronics in 2001. From 2001 to 2008, he was the President of the Ph.D. courses in electrical and information engineering at the University of Cassino, where since 2008, he has been the Director of the Ph.D. courses at the School of Engineering. He is the author or coauthor of more than 110 publications on international conferences and journals. He is primarily specialized in the field of power device modeling and characterization. His current research interests include in the fields of reliability of power devices, particularly with regards of the single event effects in radiation environment, power device failure modeling, and development of nondestructive testing apparatuses. Prof. Busatto is a member of the Italian Electric, Electronic and Telecommunication Association. In 1991, he was the Secretary of the “Materials and Devices for Power Electronics Conference.” In 2006, he was the General Chairman of the “4th International Conference on Integrated Power Systems,” and in 2010, he was the General Chairman of the “21st European Symposium Reliability of Electron Devices, Failure Physics, and Analysis.” He has been the National Coordinator of several research projects founded by the Italian Ministry of the Research and by the Italian Ministry of the Industry. He is the member of the Technical Committees of International Conference on Integrated Power Systems, European Conference on Semiconductor Reliability and Failure Analysis, International Seminar on Power Semiconductor Devices, and IEEE Energy Conversion Conference and Exposition. He is the Reviewer of the IEEE TRANSACTION ON ELECTRON DEVICES, the IEEE TRANSACTION ON POWER ELECTRONICS, the IEEE TRANSACTION ON INDUSTRY APPLICATION, the IEEE TRANSACTION ON MICROELECTRONICS RELIABILITY AND MICROELECTRONICS RELIABILITY.

Francesco Iannuzzo (M’01) received the Laurea degree in electronic engineering in 1997, and the Ph.D. degree in electronic and information engineering from the University of Naples, Naples, Italy, in 2001, where he was involved in a study on the reliability of power MOSFETs during diode operations. He is currently an Assistant Professor of digital electronics and digital communication electronics at the University of Cassino, Cassino, Italy. He is the author or coauthor of more than 50 publications on international conferences and journals. He is primarily specialized in the field of power device modeling. His research interests include in the fields of reliability of power devices, particularly against cosmic rays, power device failure modeling and development of nondestructive testing apparatuses, including modern ultrafast field-programmable gate array based control techniques. Prof. Iannuzzo is a member of the Italian Electric, Electronic, and Telecommunication Association, and in 2010, he is the Technical Programme Committee Chair for the European Symposium on Reliability of Electron Devices, Failure Physics, and Analysis. He is a Reviewer of the IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS.