Increased Bandwidth Evanescent Open-Ended ... - IEEE Xplore

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 60, NO. 11, NOVEMBER 2012

ACKNOWLEDGMENT The authors wish to thank D. Zhang for numerous discussions on the equivalent circuits and Dr. H. Ke for his helpful suggestions and discussions on the inductor model.

REFERENCES [1] T. T. Wu, “Theory of the dipole antenna and the two-wire transmission line,” J. Math. Phys., vol. 2, no. 4, pp. 550–574, Jul. 1961. [2] D. Lamensdorf, “An experimental investigation of dielectric-coated antennas,” IEEE Trans. Antennas Propag., vol. AP-15, no. 6, pp. 767–771, Nov. 1967. [3] J. H. Richmond and E. H. Newman, “Dielectric coated wire antennas,” Radio Sci., vol. 11, no. 1, pp. 13–20, Jan. 1976. [4] B. D. Popovic, A. R. Djordjevic, and N. M. Kircanski, “Simple method for analysis of dielectric-coated wire antennas,” Radio Electron. Eng., vol. 51, no. 3, pp. 141–145, Mar. 1981. [5] Z. Shen and R. H. Macphie, “Input admittance of a multilayer insulated monopole antenna,” IEEE Trans. Antennas Propag., vol. 46, no. 11, pp. 1679–1685, Nov. 1998. [6] C. Y. Ting, “Theoretical study of finite dielectric-coated cylindrical antenna,” J. Math. Phys., vol. 10, no. 3, pp. 480–493, Mar. 1969. [7] B. P. Sinda and S. A. Saoudy, “Rigorous analysis of finite length insulated antenna in air,” IEEE Trans. Antennas Propag., vol. 38, no. 8, pp. 1253–1258, Aug. 1990. [8] L. J. Chu, “Physical limitations of omni-directional antennas,” J. Appl. Phys., vol. 19, pp. 1163–1175, Dec. 1948. [9] T. G. Tang, Q. M. Tieng, and M. W. Gunn, “Equivalent circuit of a dipole antenna using frequency-independent lumped elements,” IEEE Trans. Antennas Propag., vol. 41, no. 1, pp. 100–103, Jan. 1993. [10] B. Long, P. Werner, and D. Werner, “A simple broadband dipole equivalent circuit model,” in Proc. Int. IEEE AP-S Symp., Jul. 2000, pp. 1046–1049. [11] G. W. Streable and L. W. Pearson, “A numerical study on realizable broad-band and equivalent admittances for dipole and loop antennas,” IEEE Trans. Antennas Propag., vol. AP-29, no. 5, pp. 707–717, Sep. 1981. [12] M. Hamid and R. Hamid, “Equivalent circuit of dipole antenna of arbitrary length,” IEEE Trans. Antennas Propag., vol. 45, no. 11, pp. 1695–1696, Nov. 1997. [13] Y. Liao, T. H. Hubing, and D. Su, “Equivalent circuit for dipole antennas in a lossy medium,” IEEE Trans. Antennas Propag., vol. 60, no. 8, pp. 3950–3953, Aug. 2012. [14] B. D. Popovic and A. Nesic, “Generalisation of the concept of equivalent radius of thin cylindrical antennas,” IEE Proc., vol. 131, no. 3, pt. H, pp. 153–158, Jun. 1984. [15] B. D. Popovic, M. B. Dragovic, and A. R. Djordjevic, Analysis and Synthesis of Wire Antennas. New York: Wiley, 1982, pp. 100–108. [16] J. Moore and M. A. West, “Simplified analysis of coated wire antennas and scatters,” IEE Proc. Microwaves Antennas Propag., vol. 142, no. 1, pp. 14–18, Feb. 1995. [17] J. P. Y. Lee and K. G. Balmain, “Wire antennas coated with magnetically and electrically lossy material,” Radio Sci., vol. 14, no. 3, pp. 437–445, May 1979. [18] R. W. P. King and C. W. Harrison, Antennas and Waves: A Modern Approach. Cambridge, MA: MIT Press, 1969, p. 407. [19] D. Jaisson, “Simple model for the input impedance of a wire monopole radiator with a dielectric coat,” IET Microwaves Antennas Propag., vol. 2, no. 4, pp. 316–323, May 2008. [20] S. A. Schelkunoff and H. T. Friis, Antennas: Theory and Practice. New York: Wiley, 1952, p. 306. [21] J. D. Kraus, Antennas, 2nd ed. New York: McGraw-Hill, 1988, p. 227. [22] “FEKO User’s Manual,” EM Software & Systems-S.A. (Pty) Ltd, 32 Techno Avenue, Technopark, Stellenbosch, 7600, South Africa, 2011.

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Increased Bandwidth Evanescent Open-Ended Waveguide Antenna Design Using the Imaginary Smith Chart Peter Ludlow and Vincent Fusco

Abstract—We present a novel antenna matching technique that uses the Imaginary Smith Chart to permit wideband matching of an evanescent open-ended waveguide antenna using a dielectric sheet air-spaced from the aperture. The fabricated antenna design is demonstrated to have a measured bandwidth of 24%, from 2.13–2.7 GHz, for reflection coefficient , with 2.725 GHz waveguide cutoff. The antenna’s maximum at the upper frequency in the bandwidth aperture dimension is and so it is suitable for use in a wide angle scanning phased array. Index Terms—Electrically small, evanescent waveguide antenna, Imaginary Smith Chart.

I. INTRODUCTION Phased arrays are a well-established technique to produce steerable directive beams [1]–[3]. Waveguide or cavity type radiators are particularly suitable for applications that call for medium to high power handling as well as highly isolated array elements (e.g., for wide angle scanning). Despite these favourable characteristics, waveguide and cavity radiators for practical phased array systems suffer from limited bandwidth and/or large geometrical dimensions. Evanescent open-ended waveguide radiators are formed from an open-ended waveguide that is operated below its cutoff frequency and are therefore more compact electrically than above-cutoff open-ended waveguide antennas. This feature can alleviate the limitations associated with the emergence of grating lobes in a phased array. However, the electromagnetic fields in a standard evanescent waveguide are reactive and decay exponentially; furthermore, the impedance of an unmodified evanescent waveguide aperture is complex and dispersive in nature, hence the excitation of an open-ended aperture that can effectively couple to free-space over a substantial bandwidth remains a significant challenge. Generally, open-ended waveguide antennas with aperture dimenare dielectric-filled to enable propagation [4]–[7]. Disions electric blocks of the required size/permittivity may be difficult to obtain and thus be prohibitively expensive or add mass. The design dielectric filling and an evanesof [4] uses waveguide with cent air gap at a distance from the aperture to match from the waveguide’s characteristic impedance to free space. Here an 8% bandwidth is reported, with aperture dimenfor reflection coefficient across the bandwidth. A phased array of sions circular waveguides is presented in [5], with impedance matching of section (with ) each aperture achieved through a lower placed at a certain distance from the aperture of each element, which dielectric bandwidth is reare otherwise filled with . Reference [6] proposes exported for reflection coefficient citing a dielectric-filled waveguide using an L-shaped feed, whereby ) protrudes in a conical-shape from the dielectric (which has Manuscript received October 04, 2011; revised February 13, 2012; accepted June 18, 2012. Date of publication July 10, 2012; date of current version October 26, 2012. This work was supported by the U.K. Engineering and Physical Science Research Council under Grant EP/E01707X/1. The work of P. Ludlow was sponsored by Powerwave Technologies and by the Department of Learning (DEL) for Northern Ireland. The authors are with the Institute of Electronics, Communications and Information Technology, Queen’s University Belfast, Queen’s Island, Belfast BT3 9DT, Northern Ireland, U.K. (e-mail: [email protected]). Digital Object Identifier 10.1109/TAP.2012.2207673

0018-926X/$31.00 © 2012 IEEE

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the aperture. A match over a 27% bandwidth is achieved for reflection , with aperture width/height across the coefficient bandwidth. The design technique of [7] uses partially dielectric loaded rectangular waveguides—whereby the centre of the waveguide is airfilled and allows propagation of a pseudo-TEM wave—to achieve antenna designs that have aperture dimensions equal to and at the operating frequency, for reflection coefficient bandwidths of 0.65% and 1.2%, respectively. In [8], a square evanescent waveguide aperture is matched to free space using double ring resonators printed on to an E-plane oriented dielectric substrate that is centred in the waveguide. The design exploits the principle that backward wave propagation in an evanescent waveguide is possible if the waveguide is loaded in such a way that the effective permeability in the transversal direction of the waveguide is negative. A bandwidth of 5% for reflection coefficient is reported with aperture height/width and antenna across the bandwidth. In [9], [10], square evanescent waveguide dual-polarised phased array antennas are presented that are impedance matched to free space through rear-fed coaxial probes that resonate variable width strips placed across the aperture, with a dielectric sheet air-spaced from the aperture to permit wide-angle scanning. The phased array presented over a 42% in [9] operates with reflection coefficient across the bandwidth. bandwidth, with aperture height/width In [10], 69% bandwidth for a typical reflection coefficient of is reported, however high cross-polarisation levels are present due to the asymmetric feeding arrangement used. A method was proposed in [11] by which evanescent-mode structures may be analysed using an Imaginary Smith Chart. The Imaginary Smith Chart was used in [12], [13] to design a narrowband match to the aperture admittance of an evanescent open-ended waveguide. In [12] this resulted in a narrowband antenna design with a simulated bandand a maximum rewidth of 3.2% for reflection coefficient alised gain of 3.7 dBi. Here a capacitive post and a length of evanescent waveguide were used to achieve the desired matching, with the antenna in size. In [13] a reconfigurable evanescent open-ended waveguide antenna was presented that permitted tuning of reflection coefficient at the coaxial input port from 2.05–2.49 GHz, with a maximum realised gain of 5 dBi. This is achieved through the use of a printed iris with a shunt varactor diode connected across the waveguide aperture, at 2.49 GHz. with the maximum dimension of the antenna The work we now present uses a modification of the design procedure outlined in [12], [13] and thereby enables wideband evanescent open-ended waveguide antennas to be designed. This is achieved through using a dielectric sheet air-spaced from the aperture of the waveguide—thereby creating a shunt capacitance at the waveguide aperture—and a length of evanescent waveguide. No dielectric filling of the waveguide is required and an SMA coaxial feed is used, thereby enabling simple fabrication. The antenna design is compact in both longitudinal and transverse dimensions and allows wide matching bandwidths to be obtained. The structure of the communication is as follows. In Section II the effect on waveguide aperture admittance of adding an air-spaced dielectric sheet is examined using electromagnetic simulations. The Imaginary Smith Chart is then used to derive an antenna design with a desired impedance matched bandwidth of 2.2–2.7 GHz in waveguide with 2.725 GHz cutoff. The measured input matching and farfield results are then presented in Section III and compared with simulated results. II. THEORY OF OPERATION AND DESIGN When matching an evanescent open-ended waveguide to free space it is important to know how its aperture admittance varies with frequency. The CST electromagnetic simulation software package [14]

Fig. 1. Variation in aperture admittance of waveguide with , radiating into free space with frequency.

may be used to determine how the aperture admittance changes with frequency. The CST predicted variation in the aperture admittance of mode of a waveguide with , radithe ating into free space with frequency, below the cutoff frequency of the mode (2.725 GHz for waveguide of these dimensions), is shown in Fig. 1. One method of exciting an open-ended waveguide antenna is to use a coaxial probe inserted through a side wall of the waveguide. For an effective antenna, matching to the aperture admittance (and thereby to free space), as well as a good match from the input impedance to the probe (50 ) to a real impedance along the waveguide plane at which the probe is inserted, have to be simultaneously obtained. The latter match may be obtained, with a coaxial probe, by varying the length of the probe and the distance from the probe to the short circuit back wall of the waveguide. It is therefore evident that if the impedance at a plane along a length of evanescent waveguide can be made purely real then it is possible to form a simple antenna element. To achieve this we refer to the Imaginary Smith Chart. If the aperture impedance of the evanescent waveguide can be transformed such that for a given circle of the frequency it intersects with the lower half of the Imaginary Smith Chart at a plane in the waveguide then it is possible to match to the real impedance at this plane using a coaxial probe. Note that a length of evanescent waveguide moves an impedance point radially closer to the centre of the Imaginary Smith Chart by the factor , due to the propagation constant being purely real [11]. A dielectric sheet air-spaced from the aperture is used in the evanescent waveguide antenna phased array designs of [9], [10] to enable wide-angle impedance matching (WAIM). The concept of using a WAIM dielectric sheet to equalize the reflection coefficient for E and H polarisation at any specified scan angle was proposed in [15], with the reflective properties of the high dielectric sheet being both angle of incidence and polarisation dependent. In the antenna design presented here, a dielectric sheet air-spaced from the aperture is used as part of the antenna matching process, as it effects a shunt capacitance at the aperture plane. The magnitude of this capacitance is dependent on dielectric thickness, , relative permittivity, , and the distance, , that the dielectric sheet is air spaced from the aperture. Figs. 2–4 show the variation in simulated aperture susceptance with frequency when an air-filled waveguide of the above dimensions , , waveguide wall thickness, ) ( has a dielectric sheet placed in front of the aperture, with , and varied, respectively. Note that the effect on the aperture conductance of placing a dielectric sheet in front of the aperture is negligible. Figs. 2–4 show that the resonance in the aperture susceptance may be moved down in frequency, effecting a higher value of shunt capacitance at the aperture plane, if the dielectric sheet is placed closer to the aperture, made of a material with a higher or is thicker. A design was developed using waveguide with , for a desired lower frequency in the band of 2.2 GHz. It is useful to plot the variation in the evanescent


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Fig. 2. Simulated variation in aperture susceptance with frequency for open, , . ended waveguide with dielectric sheet, Fig. 6. Locus of dielectric sheet-transformed aperture plane admittance points plotted on Imaginary Admittance Smith Chart.

Fig. 3. Simulated variation in aperture susceptance with frequency for open, , ended waveguide with dielectric sheet, . Fig. 7. Locus of dielectric sheet/length of evanescent waveguide-transformed aperture plane admittance points plotted on Imaginary Admittance Smith Chart.

Fig. 4. Simulated variation in aperture susceptance with frequency for open, , . ended waveguide with dielectric sheet,

Fig. 5. Locus of aperture admittance points plotted on Imaginary Admittance Smith Chart.

waveguide’s aperture admittance with frequency on the Imaginary Admittance Smith Chart, as shown in Fig. 5. It is evident from Fig. 5 that it is possible to match from 2.5–2.67 GHz by using only a length of evanescent waveguide to transform the aperture admittance points such circle of the Imagthat they intersect with the lower half of the inary Smith Chart (which is a locus of purely real impedance values);

Fig. 8. Cross-section of evanescent open-ended waveguide antenna matched using dielectric sheet modelled in CST.

however to match over a wider band it is necessary to first transform the aperture admittance points using a shunt capacitance. In the antenna designs of [12], [13] the waveguide aperture admittance is transformed on the Imaginary Smith Chart such that it is moved close to the short-circuit position [12] or open-circuit position [13], thereby giving a narrow matched bandwidth. In the design presented here, it is desired that the aperture admittance is transformed at the lower frequency in the band to the point on the Imaginary Smith Chart . By , i.e., where where the impedance is equal to transforming to this point it is possible to obtain a wider band match than that reported in [12], [13], as the shunt-capacitance-transformed locus of admittance points is shaped similarly to the lower half of the circle of the Imaginary Smith Chart; therefore, when a length of evanescent waveguide is added to transform the locus of points to intersect with this half-circle, it does so over a wider frequency range. Furthermore, it is easier to design a coaxial probe that will match over a wider band to impedances that lie on the Imaginary Smith Chart beand points. tween the The aperture conductance/susceptance of an evanescent open-ended waveguide is lower for frequencies further below cutoff (as shown in Figs. 1 and 5) and admittance points further below cutoff therefore have

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Fig. 9. Simulated variation in mutual coupling with array spacing for evanescent waveguide antenna matched using dielectric sheet.

Fig. 11. Variation in measured/simulated reflection coefficient of the open. ended waveguide antenna with frequency

Fig. 10. Fabricated open-ended waveguide antenna.

to be transformed further along a circle of constant conductance to inline on the Imaginary Smith Chart. This tersect with the imposes a limitation, for given waveguide transverse dimensions, on the lower frequency to be matched since to match further below waveguide cutoff a greater susceptance has to be added i.e., a larger capacitance. This means placing the dielectric sheet closer to the aperture or using a thicker/higher permittivity dielectric, which may not be practical, and determines the lower end of the matching band. , In this design a Taconic RF-60A dielectric sheet ( i.e., two 3.18 mm dielectric sheets are stacked together) is used for matching. Using CST it was found that placing at 2.2 the dielectric sheet 0.7 mm from the aperture allows GHz to be transformed to the point on the Imaginary Smith Chart . The locus of aperture plane admittance points where at this stage in the matching process is plotted using the Imaginary Admittance Smith Chart as shown in Fig. 6, for frequencies from 2–2.67 GHz. It is evident from Fig. 6 that the locus of aperture plane admittance points (which has been transformed by the dielectric sheet) traces a similar shape in the Imaginary Admittance Smith Chart to that of the susceptance circle. A 24 mm length of evaneslower-half of the cent waveguide may be used to transform the admittance locus shown at 2.2 GHz and the admittance locus in Fig. 6 such that now intersects at multiple points along the lower-half of the susceptance circle, as shown in Fig. 7. The next stage in the design process is to design the coaxial feed to the antenna—the feed probe is the inner conductor of a 50 multicomp SMA connector inserted through the centre of the broad wall of the waveguide, thereby allowing for easy fabrication. CST was used to optimize the length of the feed probe and the distance from the back wall of the waveguide, with the matching desired to give a reflection over as wide a bandwidth as possible. Wavecoefficient guide wall thickness is equal to 2 mm. To accommodate the SMA connector feeding each element in an array, the element spacing would be greater than that given by the aperture dimensions, however scanning to a wide angle could still be obtained. By feeding the antenna in the mode centre of the waveguide’s broad wall, we ensure that the is strongly excited rather than higher-order modes. Also, the matching

Fig. 12. Variation in measured/simulated radiation characteristics of the openended waveguide antenna with frequency.

process ensures that the mode is well matched at the waveguide plane where the feed is inserted, whereas higher order modes should not be well matched as this plane. The final design is shown in Fig. 8. from The simulated reflection coefficient of the antenna is 2.09–2.67 GHz i.e., a bandwidth of 24.3%. It is useful to examine in an array environment how the mutual coupling between each evanescent waveguide antenna varies with array spacing. Fig. 9 plots how across the bandwidth between two evanesthe maximum value of cent waveguide antennas varies with array spacing for a vertically or horizontally stacked array. It is evident that mutual coupling decreases more rapidly with increased array spacing for a horizontally stacked array compared with a vertically stacked array. This demonstrates that E-field coupling is significantly greater than H-field coupling for this mode being the primary mode antenna, as is expected due to the of excitation. III. MEASURED RESULTS The open-ended waveguide antenna designed in Section II was fabricated as shown in Fig. 10. Aluminium is used to form the waveguide walls. Two sheets of RF-60A dielectric (with all copper metallization removed) were adhered together using tape, along with foam of approximately 0.7 mm thickness, which is used to give the required spacing from the aperture of the waveguide. Due to the milling process used to remove the metallization from the dielectric, the thickness of the dielectric was decreased from that specified—this means that the dielectric sheets have a combined thickness of 5.79 mm rather than 6.36 mm; in addition there exists a small air-gap of uncontrolled thickness between the dielectric sheets. Fig. 11 shows the measured/simulated variation with frequency of the matching at the input port. Note that the simulated result incorporates the actual thickness of the dielectric sheets rather than the ideal design value. The antenna is matched over a measured bandwidth of 23.6%, from , i.e., below the dom2.13–2.7 GHz, for reflection coefficient inant mode cutoff frequency of the waveguide (which is 2.725 GHz).


Fig. 13. Radiation patterns of open-ended waveguide antenna measured at (a) 2.2 GHz, , (c) 2.6 GHz, plane, , (d) 2.2 GHz, , , .

DIMENSIONS

OF

TABLE I EVANESCENT OPEN-ENDED WAVEGUIDE ANTENNA [ALL DIMENSIONS IN mm]

The antenna is matched over a very similar bandwidth to that predicted by simulation of 24.3% (2.11–2.69 GHz) for reflection coeffi. The disparity is most likely due to the air gap at the cient aperture being slightly greater than 0.7 mm. The antenna is across the bandwidth in length and its aperture is of interest. The radiation patterns of the antenna were measured in the plane and the plane. The variation in the measured/simulated peak realised gain and radiation efficiency across the bandwidth is shown in Fig. 12. Fig. 12 shows that the measured realised gain varies from 1–6.1 dBi across the bandwidth, while simulated realised gain varies from 3.5–4.1 dBi. The disparity between measured and simulated realised gain results is partly accounted for by the lower measured radiation efficiency observed in Fig. 12, which varies from 77–89.5% across the bandwidth. Measured front-to-back ratio varies from 5.4–11.4 dB across the bandwidth while simulated front-to-back ratio varies from 3.4–4.4 dB. The disparity between simulated and measured results may be due to fabrication tolerances, in particular the joints visible in Fig. 10 between the sides of the waveguide and its top/bottom, which isn’t modelled in the simulations. The measured and simulated radiation patterns of the antenna are and the planes for shown in Fig. 13 in the selected frequencies in the antenna’s operating band. Note that the CST plane is less than predicted cross-polarisation in the and therefore only the measured cross-polarisation was plotted in this

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, , (e) 2.4 GHz,

, (b) 2.4 GHz, ,

, , (f) 2.6 GHz,

plane. It may be observed that cross polarisation is fairly low at boresight, with a measured cross-polar ratio in the plane in the plane observed across the bandwidth for and the patterns shown. Cross polarisation levels are however much higher plane. at the sides and back of the antenna, particularly in the plane for for the There is a null in the measured co-polarised patterns shown, which is generally deeper than that in the simulated patterns. This corresponds to the null position in the co-polarised field. While nulls in the simulated cross-polarised patplane occur at boresight, nulls in the measured tern in the and , i.e., on either cross-polarised pattern occur between side of boresight, suggesting some asymmetry in the current distributions on the fabricated antenna. IV. CONCLUSION A method has been presented that allows the Imaginary Smith Chart to be used to design wideband evanescent open-ended waveguide antennas. The complex admittance at the aperture plane is transformed—using a dielectric sheet air-spaced from the aperture and a length of evanescent waveguide—to a real admittance at a distance from the aperture, which may then be matched to 50 using an SMA coaxial probe. The fabricated antenna design achieves a measured , with 2.725 bandwidth of 24% for reflection coefficient GHz waveguide cutoff. Measured realised gain varies from 0.5–6.1 dBi across the bandwidth and the antenna’s maximum dimension is at the upper frequency in the bandwidth. ACKNOWLEDGMENT The authors would like to thank M. Major, J. Knox, and J. Megarry for the construction of the antenna elements used in this work and A. R. Lopez, the author of [9], for providing us with his report. The authors would also like to thank G. Conway for assistance in making measurements.

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REFERENCES [1] H. A. Wheeler, “The radiation resistance of an antenna in an infinite array or waveguide,” Proc. IRE, vol. 36, no. 4, pp. 478–487, Apr. 1948. [2] L. Stark, “Microwave theory of phased array antennas—A review,” Proc. IEEE, vol. 62, no. 12, pp. 1661–1701, Dec. 1974. [3] R. J. Mailloux, “Phased array theory and technology,” Proc. IEEE, vol. 70, no. 3, pp. 246–291, Mar. 1982. [4] M. Tian, D. P. Tran, and L. P. Ligthart, “Aperture admittance and matching technique of dielectric filled waveguide antennas,” presented at the Eur. Microwave Conf., Madrid, Spain, Sep. 1993. [5] J. J. Lee and R. S. Chu, “Aperture matching of a dielectric loaded circular waveguide element array,” IEEE Trans. Antennas Propag., vol. 37, no. 3, Mar. 1989. [6] R. Chernobrovkin et al., “Wide-band antenna array,” presented at the Eur. Microwave Conf., Amsterdam, Netherlands, Oct. 2008. [7] M. N. M. Kehn and P. S. Kildal, “Miniaturized rectangular hard waveguides for use in multifrequency phased arrays,” IEEE Trans. Antennas Propag., vol. 53, no. 1, pp. 100–109, Jan. 2005. [8] S. Hrabar, “Miniaturized open-ended radiator based on waveguide filled with uniaxial negative permeability metamaterial,” presented at the IEEE Antennas Propag. Society Int. Symp., Washington DC, Jul. 2005. [9] A. R. Lopez, “Wideband dual-polarised element for a phased array antenna,” Wheeler Laboratory, Hazeltine Corporation, Greenlawn, NY, 1974, Tech. Rep. No. AFAL-TR-74-000. [10] S. J. Foti and M. W. Shelley, “An experimental wideband polarisation diverse phased array,” Proc. Military Microwaves, no. 7, pp. 263–271, Jul. 1990. [11] E. D. Jersey, “Imaginary Smith Chart for evanescent-mode structures,” Electron. Lett., vol. 16, no. 3, pp. 93–94, Jan. 1980. [12] P. Ludlow and V. Fusco, “Matching evanescent open-ended waveguide antennas using the Imaginary Smith Chart,” presented at the Eur. Conf. Antennas Propag., Rome, Italy, Apr. 2011. [13] P. Ludlow and V. Fusco, “Reconfigurable small-aperture evanescent waveguide antenna,” IEEE Trans. Antennas Propag., vol. 59, no. 12, pp. 4815–4819, Dec. 2011. [14] Computer Simulation Technology (CST) Microwave Studio 2010 [Online]. Available: www.cst.com [15] E. G. Magill and H. A. Wheeler, “Wide-angle impedance matching of a planar array antenna by a dielectric sheet,” IEEE Trans. Antennas Propag., vol. AP-14, no. 1, pp. 49–53, Jan. 1966.

Low-Cost, Microstrip-Fed Printed Dipole for Prime Focus Reflector Feed Mohmamad Qudrat-E-Maula and Lotfollah Shafai

Abstract—A low cost printed dipole antenna is proposed as a feed for prime focus reflectors. Its geometry is arranged such that the two dipole arms are on opposite sides of a dielectric substrate, fed by a microstrip line. A printed dipole-reflector is placed parasitically in front of the radiating dipole arms to make its radiation backward, toward the microstrip line. This allows mounting the feed centrally on a parabolic reflector from its apex, eliminating the needs for supporting struts. The feed antenna offers an impedance bandwidth of 16.5%, when printed on a substrate of dielecat 3 tric constant 2.5, and an overall dimension of 60 60 1.58 GHz. Its impedance bandwidth is enhanced up to 29.1% by modifying the feed line. The pattern symmetry and cross-polarization of the antenna are improved by slightly changing the arms length. Its performance on a small and , is experimenavailable deep dish, with tally calculated at 3 GHz, and overall efficiency of 55% is measured. For further verification, a second feed for operation at 6 GHz is also designed and verified on the same reflector, providing the same level of efficiency. Index Terms—Microstrip fed, prime focus reflector, printed dipole antenna, reflector feed.

I. INTRODUCTION Reflector antennas are used in terrestrial and satellite communications due to their high gains and simplicity of geometry. In these antennas, the performance depends primarily on the feed, as the reflector geometry is fixed. In prime focus reflectors, the feeds are normally waveguide feed horns or coaxial waveguides [1], [2], located at the focal point of the main reflector. However, the presence of the feed assembly can increase the reflector blockage and worsen the radiation patterns and gain. The struts that support the feed can also deteriorate the side lobe and the cross-polarization levels. One possible solution to these problems is the rear-radiating feeds such as the splash plate feed [3], [4], ring feed [5], [6], cup feed [7], and hat feed [8], [9]. However, they also suffer from limitations specific to their geometry. Microstrip antennas have simpler geometries, smaller size, and light weight. Furthermore, they are also easier to integrate with electronics and are readily adaptable to hybrid and monolithic integrated circuit fabrication techniques at microwave frequencies [10]. Therefore, earlier attempts have been made to design microstrip based antennas as reflector feeds, to benefit from their salient features as indicated above [11], [12]. However, the selected designs provided broadside radiation patterns, which required conventional strut supports, with large aperture blockage. In this communication we present the design of a printed dipole feed, with a dipole-reflector for operation at 3 GHz. In the antenna design, we follow [13]–[16] except that the antenna main beam is directed backward, toward the feed transmission line, for backward radiation. Therefore, it can be mounted on a parabolic reflector, directly from its apex, Manuscript received December 02, 2011; revised April 16, 2012; accepted May 18, 2012. Date of publication July 11, 2012; date of current version October 26, 2012. The authors are with the Electrical and Computer Engineering Department, University of Manitoba, MB R3T 5V6, Canada (e-mail: [email protected]. ca). Color versions of one or more of the figures in this communication are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TAP.2012.2208170

0018-926X/$31.00 © 2012 IEEE