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Miniaturized Antenna Arrays Using Decoupling. Networks With Realistic Elements. Jörn Weber, Christian Volmer, Kurt Blau, Member, IEEE, Ralf Stephan, and ...
IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 54, NO. 6, JUNE 2006

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Miniaturized Antenna Arrays Using Decoupling Networks With Realistic Elements Jörn Weber, Christian Volmer, Kurt Blau, Member, IEEE, Ralf Stephan, and Matthias A. Hein Abstract—In ordinary antenna arrays, the spacing between ra2. In order to use antenna arrays in diators is usually chosen small mobile platforms, the overall dimensions and, hence, the radiator separation, must be reduced, which aggravates the problem of mutual coupling between radiators. Results are highly distorted beam patterns and greatly reduced radiation efficiency. We propose a method to design a decoupling and matching network for a 10, three-element antenna array with a radiator separation of which additionally allows arbitrary beam patterns assigned to each port—subject to certain constraints. An iterative process is employed to account for network losses, which are unavoidable in any realistic network implementation. Index Terms—Antenna array, decoupling and matching, miniaturization, passive network, superdirectivity.

I. INTRODUCTION N ORDER to improve the capacity and quality of today’s communication systems, so-called “smart antennas” are employed, which allow distinguishing between different paths of signal propagation in a scattering and fading environment with the aid of analog or digital beamforming [1], [2]. A -element antenna array permits the synthesis of radiation degrees of freedom by applying appatterns possessing propriate feeding currents to the antenna ports. The use of smart antennas in small mobile terminals, such as notebooks or handheld computers, is restricted solely by the lack of space. Placing individual radiators closer together aggravates the problem of mutual coupling between antenna ports. Ludwig states in [3] that tight radiator coupling in conjunction with superdirective modes of operation (also refer to [4] and [5]) results in significant gain reduction, caused by power mismatch, of the antenna array. Excitation of such modes, however, is inevitable to take full advantage of all available degrees of freedom. Simply increasing the generators’ transmit power is not an option in battery-powered devices. The power mismatch also applies in the receive direction, decreasing the antenna array’s ability to extract energy from the field [6]. The design of passive and lossless decoupling and matching networks (DMNs) to overcome these drawbacks has been treated, for instance, in [2], [7], and [8]. Yet little or no attention was paid to how the beam patterns get affected by the network. We have addressed this issue in [9] where we have demonstrated

I

Manuscript received October 11, 2006; revised February 2, 2006. This work was supported by the Deutsches Zentrum für Luft und Raumfahrt under Grant 50YB0509, and by the Network of Excellence in Wireless Communications under the European Community 6th Framework Programme. The authors are with the Institute for Information Technology, Technische Universität Ilmenau, 98684 Ilmenau, Germany (e-mail: [email protected]) Digital Object Identifier 10.1109/TMTT.2006.874874

Fig. 1. p-port antenna system Y~ consisting of the antenna admittance matrix Y~ connected to the DMN Y~ .

a method to decouple and match an antenna array with reduced radiator separation using a lossless network having predefined radiation patterns associated with each system port—subject to certain constraints that will receive continuative attention in this paper. This approach will be useful for small terminals where a switched-beam antenna array could make up for the lack of processing power that otherwise would be required for (continuous) beamforming. The practical implementation of such a network turned out to be a challenge. This is partly owed to the fact that losses are inherent in the network and cause a distortion of the network frequency response. Degradation of the decoupling and matching performance, as well as distorted radiation patterns are undesirable results. This paper puts forward an algorithm to come up with an adjusted network matrix that counteracts these effects. Section II gives an overview about the theoretical background, recapitulates the design methods used in the hypothetical lossless case, and extends this theory to include network losses. Section III provides a proof of concept by applying the algorithm to a simulated group of three dipole radiators. II. THEORY A. Basic Principles and Definitions Here, the theoretical background of the realization of a miniaturized antenna system is laid out. In contrast to [9], notation was changed from impedance to admittance domain, merely because a straightforward network topology derives from the admittance matrix directly, as described in Section III-B. Fig. 1 shows the building blocks of such an antenna system. First, there is the -elantenna admittance ement antenna array, described by the . There is further the DMN, described matrix network admittance matrix . by the Network losses are entirely represented by the real part of the admittance matrix, thus, in the case of a lossless network,

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and is purely imaginary. The antenna and DMN matrix can be connected together to form the antenna system exposing ports. system admitThe system can now be described by the tance matrix . For decoupling and matching, the system admittance matrix must be diagonal with the reference ad, where mittance on its main diagonal, i.e., is the identity matrix and is real. The radiation pattern that arises when the th port of the antenna system is driven individually shall be referred to as the th port pattern. Port patterns are a direct consequence of the currents or voltages that occur at the antenna inputs when one particular system port is excited. The concept of transfer matrices is introduced to describe how a quantity at the inputs of the antenna system is transferred to the inputs of the antenna itis defined as self. The voltage transfer matrix

Fig. 2. Flowchart of the design process for a miniaturized antenna array.

where is the voltage vector at the system ports, and is the voltage vector at the antenna ports (refer to Fig. 1). Similarly a current transfer matrix can be defined as

Thus, when the th port of the system is fed exclusively by a voltage (current), while the other ports are short circuited (open circuited), the th column of the transfer matrix contains the voltage (current) vector that is fed into the antenna. The port patterns can, therefore, be completely determined by specifying columns in the transfer matrix. It is noted that the following two types of transfer matrices can be converted to one another, provided that both the antenna matrix and the system matrix are known: and In order to calculate the admittance matrix of the system from network matrix must be split its constituent parts, the submatrices according to into four

The indices refer to the reference planes in Fig. 1. of the entire system can now be The admittance matrix expressed as

matching, but also implements predetermined port patterns is presented here. Fig. 2 illustrates the design flow. As indicated, the task can be divided into the design processes of a lossless and a lossy network. Both will be treated individually and in depth later on. Port patterns are subject to certain constraints, which will be explained in Section II-D. As a consequence, a mutual design process is required to find a combination of antenna array and port patterns that is optimal. Once this is done, it is easy to calculate the admittance matrix of a lossless DMN using (3)–(6). In practice, however, a realization of the network will never be lossless. Losses in the system distort the port patterns and degrade the decoupling and matching performance. To compensate for these effects, the idea is as follows. The lossless theory leads to a network matrix, which is purely imaginary, . When the network is built, the component imi.e., , introplementations, which derive from the imaginary part of the duce losses, which are fully described by the real part network matrix. The question arises in what way has the imaginary part of the network matrix to be adjusted to counteract the has been degradation due to the real part. Sure enough, after will also have changed. In order to arrive at the adjusted, optimum solution, an iterative approach is implemented. This final network shall be referred to as the compensated network. C. Lossless Network Design

(2)

The generation of a lossless DMN will be presented here and refers to the upper part of Fig. 2. Using (1) and (2), separating into real and imaginary parts and solving for the DMN submafor the lossless case and trices while remembering that for decoupling and matching, yields

All networks are assumed to be reciprocal throughout this paper, i.e., their network matrices are symmetric.

(3)

where

(1)

B. Network Design Flow The basic steps necessary for the design of a realistic DMN, which not only fulfills the requirements of decoupling and

(4) (5) (6)

WEBER et al.: MINIATURIZED ANTENNA ARRAYS USING DECOUPLING NETWORKS WITH REALISTIC ELEMENTS

Here,

. A new matrix is also introduced to simplify the paperwork. Note . A lossless DMN that can, therefore, be calculated from (3)–(6) once the antenna and voltage transfer matrix are known. As the network is lossless, power conservation requires the to be equal to the power transferred to the injected power . These powers are defined by antenna (7) (8) , the superscript The real part of a quantity is denoted by designates the Hermitian transpose of a matrix. Equating to gives , which has to hold for any feeding vector . Thus, it can be deduced that

(9) which shows that the voltage transfer matrix is constrained by the real part of the antenna admittance matrix. As demonstrated in the Appendix, (9) can be solved for the voltage transfer matrix with the aid of the Cholesky decomposition [10], denoted by as follows:

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crepancy between desired and realizable port patterns, there are a number of degrees of freedom that can be employed to optimize the outcome, which include the following. may be multiplied by a con• Every column in stant. This has no effect on pattern orthogonality, however, it does affect power conservation and, thus, (9). • Appropriate array elements can be found by choosing modified radiators with corresponding radiation patterns and varying their parameters (e.g., dimensions). As usual, a tradeoff has to be found between the time and effort to be spent and the nonideality of the result. E. Lossy Network Design The results of Section II-C are insofar academic as they assume a purely lossless network. This is not realistic and has to be extended. Here, the losses of the DMN are introduced to correct the element values. The set of equations of the lossy DMN can be derived analogously to Section II-C. The only differences lie in the real parts (i.e., the losses) of the submatrices of the DMN. For now, the following real parts are assumed to be known: (12) (13) (14)

(10)

(15)

, yet can be chosen arbiThe matrix is unitary trarily. Apparently, the number of transfer matrices, which results in port decoupling and matching and, thus, the number of permittable port patterns, is infinite, while at the same time, restricted to the subset described by (10).

The power consideration from Section II-D is repeated, but this time including network losses. With

D. Antenna and Port Pattern Design It was shown in a fundamental paper by Stein [6], that the port patterns of any lossless decoupled and matched antenna system are orthogonal. Thus, (9) enforces not only power conservation within the DMN, but also port pattern orthogonality. The challenge is to create port patterns that satisfy more or less stringent design goals while being restricted to orthogonal beam patterns and condition (9). Nonoverlapping beams have higher chances of meeting this requirement, although this is not a necessary condition [6]. The desired port patterns result from appropriate superposition of the single element patterns and, thus, dictate the transfer matrix for a particular antenna. For instance, the Schelkunoff polynomial method [11] can be used to determine the feeding currents in the case of a dipole array. This does not necessarily fulfill condition (9). To “enforce” this condition, a unitary matrix (11) can be derived. Substituting this into (10) in conjunction with the actual antenna yields a , which results in orthogonal port patterns and is, therefore, realizable. Depending on the dis-

and with (7) and (8), it follows that elimination of the voltage vector

, and after

(16) which is the lossy equivalent to (9). Defining the abbreviations

and solving (16) for the voltage transfer matrix pendix) leads to

(see the Ap-

(17) Again, is an arbitrary unitary matrix. Equation (17) simplifies , and to (10) in the lossless case, where . Both the matrix and the term have to be Hermitian and positive definite for a solution to exist, which is the case in our examples. The main problem with the lossy DMN lies in the fact that the of the submatrices) are unactual losses (i.e., the real parts known. They are arbitrarily complicated functions of the imag. These functions are deterinary parts, i.e.,

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Fig. 3. Port patterns of the example antenna system. The plots of the: (a) first, (b) second, and (c) third ports are normalized to a lossless isotropic radiator. The thin lines represent the lossless case and the thick lines illustrate the patterns resulting from the compensated lossy DMN with = 20. The plotted dynamic range is 25 dB.

Q

mined by the type of network realization and topology. Thus, essentially, (12)–(15) can be summarized as

and the voltage transfer matrix turns out to be

(18) Thus, if a solution network exists, it must be a fixed point of the function . Due to the unknown nature of , no attempt was made to prove the existence of a solution in general. It turned out that, in practice, however, a solution can be found using the iteration process, as described in Section III. III. EXAMPLE OF NETWORK DESIGN

The resulting port patterns are shown as thin lines in Fig. 3. The initial goal of dividing the space into three sections is met; however, the radiation zeros as prescribed by the Schelkunoff method “moved” in order to fulfill condition (9). This leads to an increased sidelobe level, especially for port pattern 2, where the sidelobes are approximately 10 dB below peak. B. Network

The feasibility of our approach shall be substantiated by means of a selected example. Its purpose is to demonstrate that our algorithm actually converges and that the decoupling and matching behavior is restored after the iteration process. A. Antenna The example operates at 2.45 GHz and shall divide the illuminated space into three sections, similar to a Butler-matrix approach [12]. An array of three dipole antennas spaced apart consisting of thin perfectly conducting rods with ideal ports attached to their feeding points is chosen and simulated in CST Microwave Studio [13] in order to get the port patterns as well as the antenna admittance matrix. The antenna terminal currents are deduced from the polynomial method by Schelkunoff [11] to realize radiation zeros at and , and , and and , respectively, which results in the desired beam patterns. These currents are used as the initial condition of the design process discussed in Section II-D. The dipoles are shortened to a total length of because this results in an optimum efficiency for an . The antenna admittance matrix is then element spacing of

mS

Now a lossless DMN can be calculated using (3)–(6). This hypothetical network is used as a starting point for the iterative can be converted to an actual process. The admittance matrix realization of the DMN in terms of capacitors and inductors. is the negative of the element Each off-diagonal element of between the two corresponding ports. To realize the diagonal elements of , a shunt element at each port is necessary. The value of the th shunt element is the sum over the th row of . In a real design process, any realization of an inductance or capacitance introduces different losses in the system. To demonstrate the feasibility of our approach, we assume that all losses can be captured by one global quality factor so the losses of all elements can be approximated by

(20) For practical implementations, it is advisable to measure and tabulate the losses corresponding to a certain element value. These lookup tables can then be employed in the design process instead of (20). A lossy admittance matrix is calculated by reversing the process outlined above. To fulfill the requirement of power conservation (16), the voltage transfer matrix has to be adjusted and the unitary matrix using (17) and substituting the new

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Fig. 4. Outline of the DMN. Thick lines correspond to individual reactances.

, as obtained from (11) in the lossless case. An adjusted of the network admittance matrix is then imaginary part calculated from (12)–(15). This process starts over by recalculating the losses using the -factor, and so on, as depicted in Fig. 2. Iteration stops when the differences between subsequent iterations have become negligible. Multiple trials have shown that convergence is actually achieved within, at most, ten iterations, even for small factors. It should, however, be pointed out that convergence greatly depends on the element realization and the network topology employed. Different element realizations and network topologies introduce different losses and influence the iteration function in (18). Tests have shown that convergence cannot generally be taken for granted. C. Results and Interpretation In this example, is chosen as 20. After convergence is achieved, the correct operation of the network is verified by a simulation in Agilent ADS [14]. The antenna is imported as a three-port -parameter file and the network elements are inserted as “Inductor with ” or “Capacitor with ,” respectively. The outline is depicted in Fig. 4. The network consists of 21 elements whose values are between 0.15–6.2 pF for the capacitors and between 0.4–22 nH for the inductors. The element values, therefore, lie within viable intervals. The complete admittance matrix of the DMN is shown for reference in (19) at the bottom of this page. 1) Realization of the DMN: Experimental work to realize such a network is in progress. Difficulties arise from the tight tolerances imposed on the element values (refer to [9]). We are currently investigating elements that can be printed using standard printed circuit board (PCB) processes. Some elements have

Fig. 5. Frequency dependence of the transverse (thick lines) and shunt (thin lines) susceptances of the 5-equivalent network of an inductor printed on PCB according to the inlay. The dash patterns refer to measurement data (——), simulation with  = 10:02 (according to manufacturer, - - -), and simulation with  = 11:92 (according to own measurements using ring resonators, - 1 - 1 -).

already been verified by measurement [15] and we are confident that the required accuracies can be achieved. The quality factors of the designed elements display a continuous behavior and lie between 20–160. As an example, Fig. 5 shows the behavior of a printed inductor, designed for 1 GHz. The ground plane below the narrow conductor has been removed to alleviate the effects of stray capacitances. The RO3010 high-frequency laminate was at used as a substrate with a dielectric constant of 10 GHz, as specified by the manufacturer.1 Simulation results of the -equivalent network (dashed lines) clearly show that the shunt susceptances (thin line) can be made zero at the design frequency, leaving a series inductance (thick line) of approximately 1.83 nH. Measurements, however, produced the plots according to the solid curves of Fig. 5. Subsequent tests with ring resonators re, which is off vealed the true dielectric constant of by almost 20%. Resimulation with the actual dielectric constant (dashed–dotted curves) displays an excellent agreement at the design frequency. Due to this discrepancy in the material parameter, it was not possible to verify the complete operation of the first network built in [15]. A new design accounting for this problem is in progress. The results will be published after careful analysis. 2) Performance of the Antenna Array: The frequency response of the antenna system is simulated and depicted in Fig. 1High-frequency

laminates, Rogers Corporation, Rogers, CT, 2006.

mS (19)

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Fig. 6. Frequency response of first port matching and coupling coefficients S (——), S (---), and S (- 1 - 1 -). The compensated frequency response (thick lines) is compared to the lossless case (thin lines, panel a) and the lossy uncompensated case (thin lines, b).

=

Fig. 7. Efficiency  P =P of the compensated DMN depending on the quality factor Q for the first (——), second (---), and third (- 1 -1 -) port driven individually.

6. Only the first three scattering parameters are plotted. The remaining ones show similar behavior and have been omitted for clarity. Fig. 6(a) compares the lossy compensated case to the lossless case. The goal of decoupling and matching is met at the deGHz in the lossless, as well as in the sign frequency of lossy case. Without a DMN, the antenna has dB, dB, and dB over the same frequency range. The overall bandwidth is fairly narrow due to the high power mismatch of superdirective operation, as explained in Section I. The most prominent difference between the two cases lies in themore than twofold increase in bandwidth,which is due to the losses in the DMN. Fig. 6(b) clearly demonstrates the beneficial effect of the compensation algorithm by showing the frequency response of the uncompensated DMN. The compensated voltage transfer matrix turns out as

=

Fig. 8. Dependence of the worst port efficiency  P =P on radiator separation d and dipole length l for a element Q factor of 20. Both parameters are normalized to the wavelength .

resulting in the port patterns shown as thick lines in Fig. 3. Although the general shape of the port patterns has been preserved by the compensation algorithm, the relative sidelobe levels of the first and third port pattern have increased by approximately 5 dB, the sidelobes of the second port pattern have decreased a little. The most obvious degradation, however, is inflicted by the network losses. Looking at the network efficiency plotted for various factors in Fig. 7, taking the average efficiency at of % dB agrees well with the loss in port pattern gain apparent from Fig. 3. The generally low efficiency is again caused by the power mismatch under highly directive operation. D. Analysis of the Effects of Antenna Design As was suggested in Section II-D, an appropriately designed antenna greatly contributes to a successful DMN realization. The optimization process used when preparing the particular example shall now be discussed in more detail. Each column of the transfer matrix may be multiplied by an without affecting (9). What does get arbitrary unit phasor

affected, however, are the network matrix , the component values required to realize the network, and more importantly, the component tolerances (refer to [9]), as well as the network efficiency. A given network matrix will be realizable only if these degrees of freedom are utilized to perform an optimization with respect to either tolerances or efficiency. If the antenna design is to be based on simulation data, it is essential that these are conducted with sufficient accuracy. The effects of radiator separation and dipole length on the DMN performance are illustrated in terms of the corresponding ) in Fig. 8 and component tolerances in efficiencies (for Fig. 9. Some correlation between the two plots is clearly visible. was chosen in the exThe element length of ample to maximize network efficiency. While the tolerances at are around 1%, they quickly relax toward larger element separations. The fact that both the tolerances and the network efficiency increase as the radiator length approaches shows that a sound radiator is fundamental to proper DMN operation. The dipole radiator, chosen solely for demonstration, is probably not the best radiator for the purpose of miniaturized

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However, is not the only solution. Consider an arbitrary uni). If is a solution, then is tary matrix ( also a solution because

Returning to the original problem (A.1), two matrices are introduced as

and

and with

unitary. Equation (A.1) can now be expressed as

Fig. 9. Dependence of the critical component tolerance on radiator separation d and dipole length l. Both parameters are normalized to the wavelength .

(A.2) Expanding the expression

antenna arrays. Radiator coupling is not only affected by separation, but also by pattern overlap, as discussed in Section II-D and more extensively in [6]. More research will, therefore, be directed toward radiators, which possess some directivity by themselves and could be accommodated to exhibit less beam overlap.

and comparing to (A.2) results in

IV. CONCLUSION A new strategy for the design of a DMN has been presented. This network counteracts gain reduction due to the strong mutual radiator coupling of a miniaturized antenna array and, at the same time, divides the illuminated space into independent sections. Losses are specifically included in the theory in order to permit a realistic design. Simulated data based on a element dipole array illustrate the beneficial effects of this approach. Current and future research is dedicated to the practical implementation of a miniaturized antenna array. To improve the practical feasibility, further investigations regarding antenna element design, as well as the network implementation will be conducted. The advantage of our approach comes into its own in small mobile terminals, where lack of space prevents the use of conventional arrays.

Decomposing once more produces

(A.3) with unitary. Resubstitution of hand side

and for the left-hand side yields APPENDIX A solution for the following general quadratic matrix equation in is derived here: Thus, (A.3) becomes (A.1) Observe that (9) presents a special case of this with . As a first step, consider a decomposition of a square Her. One mitian positive definite matrix such that possible decomposition is the complex Cholesky decomposition . The Cholesky decomposition of yields [10] denoted by , which is upper triangular and unique. a matrix

Rearranging for

gives

and

yields for the right-

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Finally, combining the unitary matrices duces the final answer

with unitary and arbitrary and definite.

and

and

to

pro-

positive

Christian Volmer was born in Düsseldorf, Germany, in 1980. He received the Dipl.-Ing. degree in electrical engineering and information technology from the Technische Universität Ilmenau, Ilmenau, Germany, in 2005, and is currently working toward the Ph.D. degree in RF and microwave techniques at the Technische Universität Ilmenau. His current research activities concentrate on the application of miniaturized adaptive antenna arrays to mobile satellite communications. He co-conducts a research project supported by the German Ministry for Education and Research.

REFERENCES [1] R. G. Vaughan and J. Bach Andersen, “Antenna diversity in mobile communications,” IEEE Trans. Veh. Technol., vol. VT-36, no. 4, pp. 149–172, Nov. 1987. [2] H. J. Chaloupka and X. Wang, “Novel approach for diversity and MIMO antennas at small mobile platforms,” in Proc. 15th IEEE Int. Pers., Indoor, Mobile Radio Commun. Symp., Barcelona, Spain, Sep. 2004, vol. 1, pp. 637–642. [3] A. C. Ludwig, “Mutual coupling, gain, and directivity of an array of two identical antennas,” IEEE Trans. Antennas Propag., vol. AP-24, no. 6, pp. 837–841, Nov. 1976. [4] M. M. Dawoud and A. P. Anderson, “Design of superdirective arrays with high radiation efficiency,” IEEE Trans. Antennas Propag., vol. AP-26, no. 6, pp. 819–823, Nov. 1978. [5] M. M. Dawoud, “Scanning properties of superdirective antenna arrays,” in 8th Int. Antennas Propag. Conf., Edinburgh, U.K., Mar. 1993, vol. 2, pp. 694–697. [6] S. Stein, “On cross coupling in multiple-beam antennas,” IEEE Trans. Antennas Propag., vol. AP-10, no. 5, pp. 548–557, Sep. 1962. [7] J. B. Andersen and H. H. Rasmussen, “Decoupling and descattering networks for antennas,” IEEE Trans. Antennas Propag., vol. AP-24, no. 6, pp. 841–846, Nov. 1976. [8] V. Riech, “Remarks on decoupling- and matching-networks for small antenna arrays,” Arch. Elektron. Uebertrag., vol. 30, no. 5, pp. 204–208, May 1976. [9] J. Weber, C. Volmer, K. Blau, R. Stephan, and M. A. Hein, “Miniaturization of antenna arrays for mobile communications,” in Proc. 35th Eur. Microw. Conf., Paris, France, Oct. 2005, pp. 1173–1176. [10] G. H. Golub and C. F. van Loan, Matrix Computations, 3rd ed. Baltimore, MD: The John Hopkins Univ. Press, 1996. [11] C. A. Balanis, Antenna Theory: Analysis and Design, 2nd ed. New York: Wiley, 1997. [12] J. P. Shelton and K. S. Kelleher, “Multiple beams from linear arrays,” IEEE Trans. Antennas Propag., vol. AP-9, no. 2, pp. 154–161, Mar. 1961. [13] CST Microwave Studio. CST Microwave, Darmstadt, Germany, 2006. [14] Agilent ADS. Agilent Technol., Palo Alto, CA, 2006. [15] C. Kutscher, “Entwurf, Aufbau und Erprobung einer miniaturisierten Gruppenantenne einschließlich Speisenetzwerk mit vorgegebener Impedanzmatrix,” Diploma thesis, Dept. RF Microw. Tech., Tech. Univ. Ilmenau, Ilmenau, Germany, 2005.

Jörn Weber was born in Quedlinburg, Germany, in 1978. He received the Dipl.-Ing. degree in electrical engineering and information technology from the Technische Universität Ilmenau, Ilmenau, Germany, in 2003, and is currently working toward the Ph.D. degree in RF and microwave techniques at the Technische Universität Ilmenau. His current research activities concentrate on the design and optimization of miniaturized antenna arrays to mobile satellite communications. He co-conducts a research project supported by the German Ministry for Education and Research.

Kurt Blau (M’00) was born in 1949. He received the Diploma and Doctoral degrees in electrical engineering, information, and measurement techniques from the Technische Hochschule Ilmenau, Ilmenau, Germany, in 1972 and 1977, respectively. In 1987, he joined the Department for Microwave Techniques, Technische Hochschule Ilmenau. Since then, he has successfully conducted and completed numerous interdisciplinary research projects concerning topics such as control circuits for phased arrays, phase-locked loops (PLLs) with YIG-tuned oscillators, satellite receiver front ends, multichannel RF front ends, and wave propagation effects in sewerage pipes. He is currently a Senior Researcher with the Department of RF and Microwave Techniques, Technische Universität Ilmenau, Ilmenau, Germany. His interests cover research and development of RF and microwave circuits and their measurements, mobile antennas, and switched-mode amplifiers.

Ralf Stephan was born in 1959. He received the Diploma and Doctoral degrees in theoretical electrical engineering from the Technische Hochschule Ilmenau, Ilmenau, Germany, in 1982 and 1987, respectively. In 1987, he joined the Department of RF and Microwave Techniques, Technische Hochschule Ilmenau. Since then, he has successfully conducted and completed numerous interdisciplinary research projects concerning topics such as integrated GaAs microwave filters and broadband noise RADAR. He is currently a Senior Researcher with the Department of RF and Microwave Techniques, Technische Universität Ilmenau, Ilmenau, Germany. His research interests concern microwave devices and measurements, antennas, and antenna arrays.

Matthias A. Hein received the Diploma and Doctoral degrees in experimental physics from the University of Wuppertal, Wuppertal, Germany, in 1987 and 1992, respectively, and the Habilitation degree from University of Wuppertal, Wuppertal, Germany, in 1998. Since 1992, he has conducted interdisciplinary research on passive superconducting microwave electronics and materials. From 1999 to 2000, he was with the University of Birmingham, Birmingham, U.K., as an Engineering and Physical Sciences Research Council (EPSRC) Senior Research Fellow. In 2002, he joined the Faculty of Electrical Engineering and Information Technology, Technische Universität Ilmenau, Ilmenau, Germany, as a Professor, where he also currently heads the Department of RF and Microwave Techniques. He has been invited to numerous international workshops and summer schools. He has authored or coauthored various monographs, reviews, and approximately 190 technical papers. He has supervised approximately 40 diploma and doctoral students. His current research interests concern novel microwave concepts and materials for various applications including wireless communications and sensor technology. Dr. Hein was the recipient of the Alan Berman Research Publication Award for his work on satellite-based navigation systems in the framework of the U.S. Navy’s High-Temperature Superconductivity Space Experiment (HTSSE) project.