Power Electronics, IEEE Transactions on

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 18, NO. 2, MARCH 2003

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Design and Experimental Results of an Input-Current-Shaper Based Electronic Ballast Antonio J. Calleja, Associate Member, IEEE, J. Marcos Alonso, Member, IEEE, Javier Ribas, Associate Member, IEEE, Emilio López Corominas, Member, IEEE, Manuel Rico-Secades, Member, IEEE, and Javier Sebastián, Member, IEEE

Abstract—This paper presents some design issues and experimental results regarding the use of the input current shaper (ICS) technique to implement high power factor electronic ballasts. The ICS is placed between main rectifier and bulk capacitor to increase the conduction angle of the main rectifier diodes up to a minimum value to obtain low current harmonics injected to the mains. Two possibilities to implement ICS-based ballast are considered in this paper: the forward-based ICS and the flyback-based ICS. Experimental results obtained from two 40-W fluorescent lamp ballasts are also presented.

presented in Section III. Section IV introduces some experimental results from a laboratory prototype for a 40 W fluorescent lamp. Section V present a comparison of the proposed ballast with the traditional boost-based solutions. Finally, some conclusions are given to summarize the paper.

Index Terms—Fluorescent lamp, high-frequency operation, input current shaper, power factor correction, single-stage ballast.

Fig. 1(a) shows the equivalent circuit of the input current shaper (ICS). The important low frequency waveforms are shown in Fig. 1(b). As can be seen in Fig. 1(a), the equivalent in series circuit of the ICS consists of a dc voltage source . When the rectified line voltage with a loss free resistor is higher than the difference between the bulk capacitor voltage and the shaper voltage , the rectifier diodes begin to conduct and input current is sinusoidal as shown in Fig. 1(b). Then, neglecting the ripple across bulk capacitor, the input current with an ICS can be expressed in the following way:

I. INTRODUCTION

T

HE development of new topologies to implement highpower-factor low-cost electronic ballast has become an important subject of researching in power electronics. Several solutions can be obtained from the literature. A first solution to implement high-power-factor converters reducing component counts is based on the integration of two stages in a single-stage converter, normally by sharing one or more switches [1]–[3]. Different examples of this technique applied to the implementation of single-stage electronic ballasts can be found in the literature [4]–[6]. A second technique is presented in [7], [8], where a charge-pump type circuit is used to provide high input power factor. In this paper, a new technique based on the input current shaper (ICS) [9], [10] is proposed to implement single-stage high-power-factor electronic ballasts. The ICS equivalent circuit consists of a dc voltage source and a loss-free resistor (LFR) placed between the full-bridge diode rectifier and the bulk capacitor. Thus, the conduction angle of rectifier diodes can be increased as required to satisfy regulations. The resonant inverter is supplied from the low-ripple voltage bus across the bulk capacitor, therefore providing low lamp current crest factor and increasing lamp life expectancy. The main advantage of the proposed ballast is the avoidance of extra controlled switches and related control circuitry, thus providing a low-cost high-powerfactor electronic ballast [11]. Section II presents the basic ICS implementations and their operation when compared with the IEC-1000-3-2 requirements. The steady-state analysis together with some design issues are Manuscript received September 5, 2000; revised December 8, 2002. Recommended by Associate Editor C. Chang. The authors are with the DIEECS—Tecnologigrave, Universidad de Oviedo, Gijón 33204, Spain (e-mail: [email protected]). Digital Object Identifier 10.1109/TPEL.2003.809335

II. INPUT CURRENT SHAPER CHARACTERISTICS IMPLEMENTATION

AND

if if if (1) In (1), represents the conduction angle of the rectifier diodes. equals , the value of is equal to and the When rectifier diodes conduct during the whole semicycle of the line frequency, and the input current is a complete sinusoidal waveform. Obviously, this is an ideal case, in which the power factor (PF) is equal to 1.0 and the total harmonic distortion (THD) is equal to 0. However, in order to satisfy the IEC-1000-3-2 requirements it is not necessary to reach this ideal case. The electronic ballasts belong to the IEC-1000-3-2 Class C, lighting equipment including dimming devices, the limits for the different input current harmonics of this regulation are shown in Table I. Using the values shown in Table I for a minimum PF of 0.9, a maximum value for the THD of 30% is obtained. These two limits are normally used to rapidly check if the ballast input current satisfy the requirements of the regulation. Fig. 2 shows the PF and the THD corresponding to the ICS current waveform as a function of the conduction angle . As can be seen in Fig. 2(b), the minimum value of the conduction angle to satisfy the requirements of the IEC-1000-3-2 is 129.1 . In order to assure that the regulation is satisfied, the different harmonic values should be calculated. Fig. 3 shows the amplitudes of the different harmonics for a conduction angle

0885-8993/03$17.00 © 2003 IEEE

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Fig. 2. ICS characteristics: (a) power factor and (b) total harmonic distortion. Fig. 1. (a) Equivalent circuit of the ICS and (b) operating waveforms.

TABLE I MAXIMUM PERMISSIBLE HARMONIC CURRENT EXPRESSED AS A PERCENTAGE OF THE INPUT CURRENT AT THE FUNDAMENTAL FREQUENCY

compared with the regulation requirements. As can be satisfy the regulation. The seen in Fig. 3 the value of power factor obtained for this minimum value of the conduction . angle is In order to allow an easy design of the ICS some interesting characteristics can be obtained. Firstly, the determination of the conduction angle is made based on the following condition [see Fig. 1(b)]:

(2) By rearranging equation (2) the conduction angle as

is obtained

(3)

Fig. 3. Harmonic content of the ICS waveform for = 129:5.

Fig. 4(a) illustrates the conduction angle as a function of the relationship between the peak input voltage , the dc voltage and the bulk capacitor voltage . This level of the ICS characteristic has been obtained by plotting (3). Using the characteristic shown in Fig. 4(a) the necessary ICS for a given conduction angle and a selected bulk voltage can be obtained. capacitor voltage Secondly, another interesting characteristic is the determination of the mean input power as a function of the conduction angle . The mean input power is obtained by integrating the instantaneous input power, this is

(4)

CALLEJA et al.: INPUT-CURRENT-SHAPER BASED ELECTRONIC BALLAST

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Fig. 5. Two possible implementations of an ICS-based ballast.

to this output [9], [10]. Another possibility to implement an ICS is shown in Fig. 5(b). In this solution a flyback converter operating in discontinuous conduction mode (DCM) is incorporated to implement the ICS. In this way the flyback input . This solution should not behaves as a resistance and be viewed as a two-stage implementation in cascade, since the flyback only handle a part of the total power managed by the ballast, typically the 50%, thus providing higher efficiency. Fig. 4. Characteristics of the ICS: (a) conduction angle and (b) normalized input power.

being the instantaneous line voltage and the instantaneous input current given by (1). The normalized mean input power can be expressed as (5) being the peak input voltage and the peak input current as shown in Fig. 1(b). Fig. 4(b) shows the normalized mean input power as a function of the conduction angle . This characteristic has been obtained by plotting (4). The ICS peak input current can be obtained from (1) as (6) Then, from (5) and (6) the necessary ICS resistance as be obtained for a given input power

III. STEADY-STATE ANALYSIS AND DESIGN ISSUES A. Forward-Type ICS The steady-state analysis of the ICS based on the forward-type converter can be performed using the equivalent circuit shown in Fig. 6(a). Assuming negligible current ripple , the following expression is through the filter inductor obtained: (8) is equal to the mean value of voltage The output voltage shown in Fig. 6(b), then (9) Using (8) in (9) and rearranging, the following expression is obtained:

can (10) where (7) (11)

Fig. 5 illustrates two different implementations of the ICS. In Fig. 5(a), a transformer is used to supply the lamp providing galvanic isolation. An auxiliary secondary winding is incorporated to implement the ICS, which is constituted by a forward-type converter with an additional delaying inductor . This additional inductor provides the ICS behavior

being the peak voltage of the square waveform across the the switching primary winding of the transformer and frequency. When operating an ICS for power factor correction the interval time is changing following the low frequency variation

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Where and represent the rms and mean values of the input current given by (1), respectively, and can be obtained by integrating (1) as

(17)

(18) Using (17) and (18) in (15) and (16), respectively, the following expressions are finally obtained: (19) (20)

Fig. 6. (a) Equivalent circuit of the forward-type ICS and (b) operating waveforms.

The capacitor C1 is used to eliminate the high-frequency switching in ICS output. As can be seen in Fig. 7, the net power is sent back to the input, whereas handled by the ICS the loss free resistor power is transferred to the output. As conclusion, most of the handled power is directly transferred to the output and the total efficiency should be higher than that obtained from the two stage arrangement. B. Efficiency Study for Forward-ICS-Type

Fig. 7.

Power diagram of the forward-type ICS efficiency study.

of the input current. In order to assure a correct operation of the ICS the following condition should be satisfied: (12) by rearranging (12) and using (11), the condition can be expressed as (13) Fig. 7 illustrates the power diagram of the forward-type ICS consists of the ballast. The power processed by the ICS and the loss free equivalent the dc voltage source power , this is resistor power (14) These two power components can be calculated as

The Fig. 7 illustrates the power diagram of the forward-type ICS efficiency study. This study has been realized as a function on the handled power for each stage. The following expressions have been defined. : Is the handled power for both, shaper voltage and equivalent loss free resistor. : Is the processed power for the inverter stage. This power is directly sent to the resonant tank. : Is the power handled by the ICS. This power is sent to the input. : The power delivered to the lamp. k: Is the shared power factor: inverter power transferred to the ICS versus the power directly transferred to the output. : Power distributed handled by the ICS. This power is sent to the input. : Inverter stage efficiency. : Input current shaper efficiency. : Resonant tank efficiency. The handled power by the inverter stage is given by (21) The handled power by the shaper stage is given (22) The next expression is the directly power transferred to the output (23)

(15) (16)

The power handled by the shaper is divided in two parts. That , handled by the loss-free resistor which is sent is the say.


to the output and output

handled by the ICS which is sent to the (24)

The power sent to the input can be expressed as a function of the distributed power and the Shaper efficiency as (25) The value of the power sent to the output is given by

(26) The Lamp power delivered is

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C. Flyback-Type ICS It is well known that for a flyback converter operating in discontinuous conduction mode the mean input current is given by [12] (32) being the duty-cycle, the switching frequency, the input inductance and the instantaneous input voltage. Assuming constant duty cycle and switching frequency, the input current is proportional to the input voltage. Thus, the converter behaves as a resistive load for the line. The value of this load is given by (33)

(27)

The condition to assure discontinuous operation mode for the flyback converter is given by [12]

The total efficiency of the conversion is expressed for (34)

(28) By using (25) in (28) the next expression is obtained (29) Using this expression can be obtained the total efficiency as a function of both, distributed power and the stages efficiency

(30) In the Fig. 7 show the example of the efficiency Forward-ICS topology. The values of the efficiency by stage and the constants has been obtained of the design prototype. This values are ; ; ; ; . The efficiency of the proposed topology using (30) is

being the turn ratio of the coupled inductors, the flyback the dc output voltage existing across peak input voltage and bulk capacitor. In this case, for the flyback-based ICS, the value of the equivalent dc voltage source is equal to zero, this is (35) As can be deduced from (33), by varying the duty cycle can of the flyback converter the equivalent resistance be modified, thus controlling the output power at constant switching frequency. This results very interesting to implement high-power-factor electronic ballast with dimming feature. Fig. 8 shows an equivalent circuit of the proposed ballast. The input of the flyback-based ICS has been modeled by means of . The value of this resistor is its equivalent loss-free resistor given by (33) and can be controlled by using the duty cycle. The power theoretically dissipated by this resistance is transferred to the output current source . The instantaneous input power of the flyback-based ICS is given by (36) is given by (1). The mean input power of the flyback where ICS can be obtained as

The efficiency of the conventional topology based in two stage is the product of the converter PFC efficiency, of the inverter efficiency and the resonant tank efficiency

(37)

(31)

The mean output current of the source can then be obtained by equaling input and output power of the flyback ICS. This gives

Therefore, an increase close to five points in total efficiency in achieved.

(38)

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Fig. 9. Power diagram of the proposed ballast. Fig. 8. Equivalent circuit of the flyback-type ICS ballast.

On the other hand, the power delivered to the resonant inverter at steady state operation is given by (39) being the equivalent input resistance of the resonant inverter at steady-state operation. This resistance depends on the reactive elements, load and frequency operation of the resonant inverter [5], [6]. Fig. 9 shows the power diagram of the proposed ballast. As can be seen in this figure only part of the total input power is processed by the two stages, whereas there exists a partial power flow directly from the input to the output through the resonant inverter. The reason is that when the switch of the flyback ICS is on, the input power goes to both ICS and resonant inverter. However, when the switch is off the resonant inverter takes the power from the output of the flyback and bulk capacitor, and this power is then processed twice. The distribution of the power is then directly related to the flyback duty cycle. The total efficiency will be also higher than that obtained for the two-stage ballast.

Now we can estimate the total efficiency in this topology. If , the efficiency of the each stage are: The efficiency of Flyback-ICS-Type is (45) We can calculate the efficiency of the topology based in two . The energy is twice processed stage (46) Therefore, an increase close to 5 points in total efficiency in achieved. E. Resonant Inverter To obtain the characteristics of the inverter an analysis based on the fundamental approximation has been made. The following relationships are obtained for lamp voltage and resonant current:Per unit lamp voltage (47) Per unit inductor current

D. Efficiency Study for Flyback-ICS-Type

(48)

The input power can be calculated that (40) is the power delivered to the ICS, and is the input where inverter power. The processing power by the ICS can be expressed as (41) where k is the distributed power parameter, typically 0.4 Hence the energy processed by the inverter stage can be expressed as (42) A balance of power can be made as a function of both effi(inverter efficiency), and (Shaper efficiency). The ciency, power delivered to the lamp can be calculated as (43) The total efficiency of the conversion is given by (44)

Inductor current phase (49) being and Q the normalized frequency and load, respectively. The following base values have been used: (50) Another interesting characteristic is obtained by equaling (47) to zero. This gives the following relationship between the normalized load and the normalized switching frequency (51) This condition provides the operating points in which the resonant current is in phase with the resonant tank input voltage. At these operating points the bridge switches turn on and off in the resonant current zero crossovers, thus providing zero switching losses and avoiding reactive power handling. Equation (51) has


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been plotted in Fig. 11. As can be seen only operating points and are possible. with By using (51) in (47) and (48), the per unit output voltage and resonant current at these operating points with zero resonant current phase are obtained (52) Since input voltage and resonant current are in phase at these operating points, the inverter input power will be given by the product between the per unit rms values, this is (53)

Fig. 10. Operating points for zero resonant current phase.

Using (48) in (51) to obtain the actual power and equaling to (39), the equivalent input resistance of the resonant inverter is obtained for these operating points with voltage and current in phase (54)

IV. DESIGN EXAMPLES Two prototypes based on the ICS circuit have been built and tested at the laboratory. The corresponding design processes will be presented as an example. The lamp used was a ring-shaped fluorescent lamp TLE-40 W from Philips. This lamp was aged for 100 h and tested at high frequency obtaining the following measurements: V

V

I

The line voltage is is 50 Hz.

A

R

Ohm

Fig. 11. Electric diagram of the simulated and laboratory prototypes: (a) forward-type ICS ballast and (b) flyback-type ICS ballast.

and line frequency quency is 80 kHz. Then, using (11) the following values are obtained:

A. Forward-Based ICS Prototype The ICS is designed to operate with a conduction angle , then from Fig. 4(a) a value of is obtained for the ICS voltage ratio. The voltage across bulk capacitor is selected . to be equal to the peak line voltage, this gives can be obtained as Then, the ICS dc voltage

(57) (58)

is obFrom Fig. 4(b) a per unit input power . Then, using (7) the tained for a conduction angle equivalent resistance of the ICS is attained

Regarding the resonant tank, the values of the transformer and resonant elements and should be obturn ratio tained. In order to operate with zero phase for the resonant curand has been rent an operation point with selected from Fig. 10. From this operation point the resonant frequency and the base impedance of the resonant tank are obtained as

(56)

(59)

has been considered, assuming a 100% where a efficiency for the ballast. At this point, the turn ratio for the ICS transformer winding can be calculated. The transistor bridge and the inductance will operate with an input voltage of 310 V, thus producing a across the transformer primary square wave of winding, with a 50% duty cycle. The selected switching fre-

(60)

(55)

The resonant elements are then easily calculated (61) (62)

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For the selected operating point, by using (52) a per unit lamp is obtained. The necessary turn ratio voltage for the transformer is obtained from the following equation:

The resonant elements are then easily obtained (75) (76)

(63) Solving (63) for

the following value is obtained:

V. COMPARISON WITH BOOST-BASED SOLUTION (64)

B. Flyback-Based ICS Prototype The ICS is also designed to operate with a conduction angle with and . The voltage across bulk capacitor and the equivalent ICS resistance are obtained as (65) (66) A switching frequency of 25 kHz and nominal duty cycle of 0.5 have been selected. Then, from (33) the input inductance of the flyback ICS is calculated (67) Now, from the condition of operation in discontinuous conduction mode, the turn ratio of the flyback coupled inductors is obtained (68) has been selected to provide a safety margin. a value of Regarding the resonant tank, the fundamental component of the resonant tank input voltage will be equal to the following value: (69) The per unit output voltage is then obtained by using the rms lamp voltage (70) Since the design is performed to obtain resonant current and voltage in phase, the normalized load of the resonant tank is obtained from (52) (71) Then from (44) the necessary normalized frequency is obtained (72) The resonant frequency and base impedance of the resonant tank can now be calculated as (73) (74)

The typical solution when implementing high power factor electronic ballasts for fluorescent lamps is the use of the boost converter as pre-regulator stage. The boost converter can be operated in either discontinuous conduction mode (DCM) as in [4],[13],[14] or in the borderline between continuous and discontinuous conduction mode as in [15]. The use of the boost converter as pre-regulator operating in continuous conduction mode (CCM) highly complicate the control and sensing circuitry and therefore is only used for the higher power range. Note that for fluorescent lamps the higher lamp power is usually 60 W. When using the boost converter as pre-regulator there are also two possibilities. One is to implement the complete boost converter as an upstream stage, normally using an especial integrated circuit for driving the switch and to perform the sensing, protection and regulation functions [15]. This solution increases the cost compared to the use of the ICS, since the ICS operates automatically without using any control circuitry. Also, the use of the ICS allows to reduce the number of power switches because only two switches are required. From the point of view of the reactive elements, the inductor in the boost converter is handling a peak current of twice the maximum line current (when operated in critical conduction mode). However, in the ICS the inductor in series with the input operates always in CCM, what means that the peak current is nearly equal to the line current. On the other hand, in the boost solution operating with border line control a high switching frequency range is required (typically 30 kHz–140 kHz), whereas in the ICS the switching frequency is constant. Therefore, ICS inductor is expected to be between three and four times smaller is size. It is also true that for the ICS is necessary an additional delaying inductor, but the mean current through this inductor is low compared to the input current, what allows to minimize its volume. Besides, this inductor can be partially or even completely implemented by means of the leakage inductor of the output transformer. Regarding the bulk capacitor, in the boost solution the ripple current through this capacitor is lower than in the ICS, however the rated voltage is higher. Therefore, the size of the bulk capacitor will be similar in both solutions. Finally, the EMI filter is simplified when using ICS since it operates at constant frequency. Boost converter operated with borderline control generates electromagnetic interference in a very broad range due to switching frequency excursion. The size of common mode and differential mode chokes can be reduced in a factor between two and three. The other possibility more competitive with the ICS solution is the use of the boost converter integrated with the half-bridge inverter [4],[13],[14] In this case, only two switches are necessary as in the ICS ballast. However, the first difference is that the switch shared by the boost converter and the half-bridge inverter is forced to handle both input current and resonant current. As a


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Fig. 13. Simulated results of the flyback-type ICS: (a) bulk capacitor voltage and (b) main input voltage and current obtained from the simulation.

Fig. 12. Experimental results for the forward-type ICS: (a) main current and voltage (100 V/DIV, 0.2 A/DIV, 5 ms/DIV) (b) lamp voltage and current (50 V/DIV, 1 A/DIV, 5 us/DIV).

consequence, this switch presents a higher current stress than the other one and must be oversized. For the ICS solution the current can be equally distributed between the two switches by using a capacitor in series with the primary winding, thus allowing to use similar devices with equal characteristics and avoiding the rising of hot spots in the converter. On the other hand, the integration of the boost converter with the half-bridge inverter prevents from using the borderline control, since the frequency variation would generate a lamp power variation. Therefore, the boost converter is operated in DCM, what means a high voltage stress across the bulk capacitor and power switches, typically twice the input voltage to have a nearly sinusoidal input current [4]. In the case of the ICS solution the voltage stress is equal or even lower than the peak line voltage. This means smaller and low cost bulk capacitor and switches. Regarding the inductor, the peak current in the DCM boost converter can be as high as four times the maximum line current in a typical design, whereas

in the ICS converter is nearly equal to the input current as stated previously. Consequently, the ICS inductor will be much smaller in size and lower in cost. EMI filter chokes are also large in volume when using the DCM boost converter due to the high death time in the input current. As summary, the ICS solution is a feasible possibility when compared with the typical solutions used nowadays in commercial electronic ballasts. The main disadvantage of the ICS circuit is that the use of an output transformer is mandatory. This output transformer is normally small in size for the power range considered (for European mains and up to 60 W for fluorescent lamps, an E20 or E25 can be used). Therefore, the typical use of the ICS based ballast would be in appliances in which the use of an output transformed is required. For example in lightings for writing desks where the galvanic isolation is recommended. Another example could be in applications where the lamp voltage and line voltage are quite different and it is necessary the use of a transformer to adapt voltage levels. This is for example the case of some types of UV lamps. VI. EXPERIMENTAL RESULTS Fig. 11 shows the electrical diagrams of the two laboratory prototypes, implemented with the forward-type ICS [Fig. 11(a)] and with the flyback-type ICS [Fig. 11(b)].

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voltage and current, similar results as those obtained from simulation can be seen in this figure. Fig. 14(b) shows the resonant tank input waveforms, as can be seen operation with zero resonant current phase is achieved, thus minimizing switching and conduction losses. Measured efficiency for this prototype was 90%. VII. CONCLUSION In this paper a new solution to implement single-stage high-power-factor electronic ballasts based on the input current shaper (ICS) has been presented and evaluated. The ICS is implemented between main rectifier and bulk capacitor, thus increasing the conduction angle of the main rectifier diodes up to a minimum value to obtain low current harmonic injected to the mains. The main idea is to obtain an input current waveform good enough to satisfy standard requirements, thus minimizing the energy processed by several stages and increasing efficiency. Two possible implementations based on forward-type ICS and flyback-type ICS have been presented and analyzed. The forward-type ICS presents the advantage of avoiding extra controlled switching, thus minimizing control circuitry. However, several extra passive elements are needed. The flyback-type ICS requires the inclusion of an extra controlled switch but it also allows to easily implement a dimming feature by controlling the duty cycle of this extra switch. The proposed ballasts have been both simulated and implemented at the laboratory showing good possibilities. REFERENCES

Fig. 14. Experimental results for the flyback-type ICS: (a) main current and voltage (100 V/DIV, 0.2 A/DIV, 5 ms/DIV) (b) resonant tank waveforms: input voltage (100 V/DIV) and input current (1 A/DIV). Horizontal scale: 10 s/DIV.

Fig. 12 illustrates the input and output waveforms of the forward-type ICS ballast. Fig. 12(a) shows the line voltage and current, the measured input power factor was 0.96 and the harmonic contents of the input current is within IEC-1000-3-2 requirements. Fig. 12(b) illustrates the lamp voltage and current for an output current of about 40 W. The measured efficiency was 84%. Fig. 13 illustrates the simulation results for the flyback-type ICS ballasts. Fig. 13(a) shows the bulk capacitor voltage, which reaches the calculated value of about 138 V at steady-state operation. Fig. 13(b) shows the input voltage and current, as can be seen the conduction angle of the input rectifier is equal to the necessary value of 130 (7.2 ms for 50 Hz). Finally, Fig. 14 illustrates some experimental results obtained from the flyback-type ICS ballast. Fig. 14(a) shows the main

[1] IEC 1000-3-2 (1995-03) Standards on Electromagnetic Compatibility (EMC), Part 3, Section 2: Limits for Harmonic Current Emissions, International Electrotechnical Commission, Geneva, Switzerland, Apr. 1995. [2] M. Madigan, R. Erickson, and E. Ismail, “Integrated high quality rectifier regulators,” in Proc. IEEE PESC’92 Conf., 1992, pp. 1043–1051. [3] T. F Wu and T. H. Yu, “Off-line applications with single-stage converters,” IEEE Trans. Ind. Electron., vol. 44, pp. 638–647, Oct. 1997. [4] C. Blanco and J. M. Alonso et al., “A single-stage fluorescent lamp ballast with high power factor,” in Proc. IEEE APEC’96 Conf., 1996, pp. 616–621. [5] J. M. Alonso and A. J. Calleja et al., “Analysis and experimental results of a single-stage high-power-factor electronic ballast based on flyback converter,” in Proc. IEEE APEC’98 Conf., 1998, pp. 1142–1148. [6] , “Single-stage constant-wattage high-power-factor electronic ballast with dimming capability,” in Proc. IEEE PESC’98 Conf., Fukuoka, Japan, 1998, pp. 1330–1336. [7] M. A. Có, D. S. L. Simonetti, and J. L. F. Vieira, “High-power-factor electronic ballast operating in critical conduction mode,” IEEE Trans. Power Electron., vol. 13, pp. 93–101, Jan. 1998. [8] J. Quian, F. C. Lee, and T. Yamauchi, “Current-source charge-pump power-factor-correction electronic ballast,” IEEE Trans. Power Electron., vol. 13, pp. 564–572, May 1998. [9] J. Sebastián and M. M. Hernando et al., “Input current shaper based on the series connection of a voltage source and a loss-free resistor,” in Proc. IEEE APEC’98 Conf., Anaheim, CA, 1998, pp. 461–467. [10] , “A new input current shaping technique using converters operating in continuous conduction mode,” in Proc. IEEE PESC’98 Conf., Fukuoka, Japan, 1998, pp. 1330–1336. [11] J. M. Alonso, A. J. Calleja, J. Ribas, E. López, and M. Rico, “Using input current shaper in the implementation of high-power-factor electronic ballasts,” in Proc. IEEE APEC’99 Conf., Dallas, TX, 1999, pp. 746–752. [12] J. M. Alonso, P. J. Villegas, J. Díaz, C. Blanco, and M. Rico, “A microcontroller-based emergency ballast for fluorescent lamps,” IEEE Trans. Ind. Electron., vol. 44, pp. 207–216, Apr. 1997.


[13] R. Oliveira and J. L. F. Vieira, “High-power-factor electronic ballast with constant dc-link voltage,” IEEE Trans. Power Electron., vol. 13, pp. 1030–1036, Nov. 1997. [14] T. F. Wu, M. C. Chiang, and E. B. Chang, “Analysis and design of a high power factor single-stage electronic ballast wit dimming features,” in Proc. IEEE APEC97 Conf., Nov. 1997, pp. 1030–1037. [15] M. Bairanzade, Electronic Lamp Ballast, Motorola Semiconductor application note AN1543, 1995.

Antonio J. Calleja (S’98–A’02) Was bon in Leon, Spain, in 1964. He received the M.Sc. and Ph.D. degrees in electrical engineering from the University of Oviedo, Spain, in 1995 and 2000, respectively. From 1995 to 2002 he was an Assistant Professor at the Electrical and Electronic Department, University of Oviedo, where since 2002 he has been an Associate Professor. His research interests include high-frequency electronic ballasts, discharge lamp modeling, power factor correction topologies, high frequency switching converters, power converters for electrostatic applications, and industrial control systems. Dr. Calleja is a TRANSACTIONS Paper Reviewer and a member of the International Ozone Association (IOA).

J. Marcos Alonso (S’94–A’95–M’98) received the M.Sc. and Ph.D. degrees in electrical engineering from the University of Oviedo, Spain, in 1990 and 1994, respectively. From 1990 to 1999, he was Assistant Professor at the Electrical and Electronic Department, University of Oviedo, where since 1999 he has been an Associate Professor. He is the primary author of more than 30 journal and international conference papers in power and industrial electronics. His research interests include high-frequency electronic ballasts, discharge lamp modeling, power factor correction topologies, high frequency switching converters, power converters for electrostatic applications, and industrial control systems. He currently holds two Spanish patents and has four pending. Dr. Alonso received the IEEE Industrial Electronics Society Meritorious Paper Award for 1996. He is a TRANSACTIONS Paper Reviewer, Conference Session Chairman, and a member of the International Ozone Association (IOA).

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Javier Ribas (S‘97–A‘01) was born in Milwaukee, WI, in 1971. He received the M.Sc. and Ph.D. degrees in electrical engineering from the University of Oviedo, Spain, in 1995 and 2001, respectively. Since 1996, he has been with the Electrical and Electronic Department, University of Oviedo, where he is currently an Assistant Professor. His research interests are electronic lighting system, switching-mode power supplies, and high-power-factor rectifiers.

Emilio López Corominas (M‘97) was born in Oviedo, Spain, in 1965. He received the M.Sc. and Ph.D. degrees in electrical engineering from the University of Oviedo, Spain, in 1992 and 1999, respectively. In 1993, he joined the Electrical and Electronic Department, University of Oviedo, where he is currently Associate Professor. His research interests include high-frequency electronic ballasts, discharge lamp modeling, high-frequency switching converters, power factor correction converters, and industrial control systems.

Manuel Rico-Secades (M‘87) was born in Oviedo, Spain, in 1961. He received the M.Sc. and Ph.D. degrees in electrical engineering from the University of Oviedo, Spain, in 1986 and 1989, respectively. Since 1986, he has been with the Electrical and Electronic Department, University of Oviedo, where he is currently a Full Professor. His research interests include industrial electronics and power electronics, especially resonant converters, electronics ballast, discharge lamp modeling, dc-to-dc converters, power factor correction topologies, and industrial control. Dr. Rico-Secades received the IEEE Industrial Electronics Society Meritorious Paper Award in 1996. He is a member of the Illuminating Engineering Society of North America (IESNA).

Javier Sebastián (M’88) was born in Madrid, Spain, in 1958. He received the M. Sc. degree from the Polytechnic University of Madrid, Spain, in 1981 and the Ph.D. degree from the University of Oviedo, Spain, in 1985. He was an Assistant Professor and Associate Professor at both the Polytechnic University of Madrid and the University of Oviedo, Spain. Since 1992, he has been with the University of Oviedo, where he is currently a Professor. His research interests are switching-mode power supplies, resonant power conversion, converter modeling, and high power factor rectifiers.