technology with feedback-loop control for capacitive ... - IEEE Xplore

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IEEE TRANSACTIONS ON ELECTRON DEVICES, VOL. 47, NO. 11, NOVEMBER 2000

“Burst” Technology with Feedback-Loop Control for Capacitive Detection and Electrostatic Excitation of Resonant Silicon Sensors Thierry Corman, Kjell Norén, Peter Enoksson, Jessica Melin, and Göran Stemme

Abstract—A method for excitation and detection of resonant silicon sensors based on discontinuous, “burst” excitation is presented. The solution eliminates the crosstalk between electrostatic excitation and capacitive detection by separating them in time. High excitation voltages can be combined with highly sensitive detection electronics. The method facilitates the use of large distances between the resonator and electrodes used for excitation and detection. The method was successfully tested with feedback-loop control on silicon resonant density and pressure sensors where the electrodes were positioned outside a glass. Continuous measurements of gas pressures and liquid densities were realized. The simplified fabrication process utilized reduces the risk of leakage from the ambient pressure to the low-pressure cavities in which the resonators are encapsulated since electrical feedthroughs are not needed. Excitation voltages alternating between 0 and 150 V could be applied to the resonators with measured electronics sensitivities of 0.4 fF. Signal-to-noise ratios (SNRs) as high as 100 (density sensor) and 360 (pressure sensor) were obtained. The electronic evaluation revealed that the “burst” duty cycle (i.e., the excitation time relative to the free oscillation time) had a strong influence on the output detection voltage. As few as two excitation periods with a “burst” cycle frequency of 115 Hz and a “burst” duty cycle of 1% was sufficient to select and lock the resonance frequency (28 042 Hz) for the tested pressure sensor. The same electrodes could be used for both excitation and detection. A novel solution is also presented that eliminates the charging effect of dielectric surfaces which otherwise can be a problem for capacitive detection. Index Terms—“Burst” electronics, capacitive detection, discontinuous excitation, electrostatic excitation, feedback control, switching.

I. INTRODUCTION

E

LECTROSTATIC excitation and capacitive detection techniques are essential components for fully functional silicon resonant sensors. Excitation enables the resonator to vibrate and the generated oscillations are detected by an appropriate detection technique. In a resonance-based microsensor system, crosstalk between excitation and detection is a considerable limitation. This crosstalk makes it difficult to separate the detection signal from the excitation noise; therefore, excitation voltages are usually limited to low voltages. For electrostatic excitation and capacitive detection, the stray capacitances are

Manuscript received November 2, 1999; revised June 27, 2000. The review of this paper was arranged by Editor K. Najafi. The authors are with the Department of Signals, Sensors and Systems, Royal Institute of Technology, SE-100 44 Stockholm, Sweden (e-mail: [email protected]). Publisher Item Identifier S 0018-9383(00)09620-9.

generally of the same order of magnitude as the capacitance one wants to measure. To overcome this problem, one solution is to integrate electronic circuitry on-chip. However, this is not always possible and complicates the fabrication process. Other solutions have been proposed to separate excitation from detection in frequency using a high frequency detection signal which is then demodulated [1]. A solution was proposed where a simple resistor combined with a dc voltage generates an autonomous oscillator without needing electrostatic ac excitation [2]. Advanced electronic circuitry and/or physical shielding systems have also been proposed to minimize the parasitic capacitances [3]–[5] but remain very sensitive to external perturbations such as high excitation voltages. To achieve a high Q-factor, the mechanical resonator is often encapsulated in a low pressure cavity. Therefore, conventional systems using electrostatic excitation and capacitive detection require electrical feedthrough conductors which connect the external bonding pads to the electrodes in the low-pressure cavity. However, the use of feedthroughs has some limitations. The fabrication process of the devices becomes complicated due to sensitive steps such as anodic bonding with an intermediate insulating layer (e.g., silicon dioxide) [4] and the leakage risks of such encapsulated devices become relatively high. The small electrode-resonator distances ( 10 m) required for sufficient excitation and detection efficiency is also a problem due to squeeze-film damping losses and a consequent decrease of the quality factor. We introduce a “burst” method for excitation and detection of silicon resonators to overcome the problems of crosstalk between excitation and detection and to eliminate the need for electrical feedthrough conductors for electrostatically excited and capacitively detected resonators. A novel solution is also presented to eliminate the negative charging effect of a dielectric surface during capacitive detection. The “burst” electronics with feedback-loop control is evaluated and tested with electrostatic excitation and capacitive detection, but can be applied to other types of excitation and detection principles, i.e., piezoresistive, piezoelectric, optical, or thermal principles. This technology is applied to our earlier presented resonant fluid density sensor [6] and resonant pressure sensor [7]. We chose to evaluate the “burst” technology with two different sensors to show that it can be easily applied to other resonant sensors having different characteristics (i.e., different resonance frequencies, different Q-factors, etc.) without major design changes. The “burst” technology can be applied to both surface and bulk micromachined devices.

0018–9383/00$10.00 © 2000 IEEE

CORMAN et al.: “BURST” TECHNOLOGY WITH FEEDBACK-LOOP CONTROL

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Fig. 1. Schematic “burst” technology principle.

Fig. 3. Schematic layout of the electrodes and cross section of the tested resonant fluid density sensor. Electrodes C and C were used for electrostatic excitation (antiphase torsion mode) and electrodes C and C were used for capacitive detection.

to be selected, which can be distinctly different from the resonance frequency of the dominant mode that one would automatically select when just a pulse excitation technique is used [8]. To find the desired resonance frequency, the oscillator frequency is ramped up or down from a defined initial frequency , until resonance is found and locked by the detection circuit electronics. III. ELECTROSTATIC CHARGING EFFECT Fig. 2. Illustration of the excitation and detection time separation (not to scale). Signal A corresponds to the signal just before the detection switch.

II. PRINCIPLE OF OPERATION The principle illustrated in Figs. 1 and 2 can be described as follows. The resonator is excited during a certain time interval by an oscillating signal at the resonance frequency of the device. The excitation is then switched off and the detection starts on the now freely oscillating resonator. In this way, the excitation and detection are separated in time and do not interfere with each other. If necessary, a short time delay is introduced between the excitation time and the detection time, see Fig. 2. This delay allows the excited resonator to settle into an oscillating state without direct influence of the excitation signal. After the detection period, the excitation is once again switched on and the cycle is thus completed. The excitation is locked to the detected signal, thus keeping the resonator at resonance. Note that if the oscillation of the resonator damps out too quickly after the excitation time, i.e., if the resonator Q-factor is too low, the vibrations cannot be detected. In the “burst” technology solution, the excitation consists of a “burst” of pulses. This allows a specific resonance frequency

For capacitive microsensors and microactuators with a dielectric medium such as Si N or SiO (generally used for short-circuit protection) placed on one of the electrodes and between the two electrodes, the surface of the dielectric medium may be subjected to localized charge storage [9]. This effect can be explained as follows. When a discharge occurs in the air gap between one electrode and the dielectric surface, an avalanche of free electrons and ions is generated. They traverse the gap and are stopped and neutralized by charge exchange at the dielectric surface instead of reaching the underlying electrode. This charging phenomenon can reduce the device reliability and make detection difficult to be realized [9]. This phenomenon was observed during capacitive detection for our tested silicon resonators, which are schematically represented in Figs. 3 and 4. A glass dielectric medium was placed between the silicon movable electrode and the fixed metal electrodes. During our initial tests, no detection signal could be observed. This might be the result of the surface charge accumulation on the glass surface, which is probably due to discharged gas between the resonator and the glass medium situated beneath the metal electrodes. As a result, the electrostatic field in the gap was locally reduced, the detection voltage decreased and the excitation voltage applied to the resonator was not sufficient

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Fig. 5.

Photo of the discrete electronic prototype.

A short delay of time ( ) needed for signal stabilization is introduced directly following the excitation period. The detection period starts after the short delay, as shown in Fig. 2. The duration of the detection period is defined as . When excitation is switched ON, a phase-compensation adjustment can be used to get the new exciting signal in phase with the resonator oscillation, see Fig. 6. B. Excitation

Fig. 4. Schematic layout of the electrodes and cross-section of the tested resonant pressure sensor. Electrodes C and C were used for electrostatic excitation and electrodes C and C (or C and C ) were used for capacitive detection.

to generate oscillation. To eliminate this negative charging effect at the dielectric surface, we present a solution consisting of alternately switching the polarity of the detection bias voltage. Thereby, the accumulation of charges at the dielectric surface is eliminated by discharge of the capacitance formed between the resonator and the metal electrodes. IV. ELECTRONIC CIRCUIT An electronic circuit prototype was built on a printed circuit board. A photograph of the density sensor connected to the complete circuit prototype is shown in Fig. 5. The supply voltage of the circuit was 15 V and the implemented electronics can be described with the help of the block diagram shown in Fig. 6. Note that the supply voltage of 15 V is amplified to 150 V in the electronics circuit for excitation and detection purposes. A. Switching Principle To separate the excitation from the detection in time, an oscillator (“oscillator 1” in Fig. 6) running at a fixed frequency is connected to two separate switches, namely switch 1 and switch 2, controlling the excitation and the detection, respectively. These two switches work in antiphase, i.e., when one is in position ON, the other is in position OFF, and vice-versa. (while the detecThus, excitation is applied during a time tion switch 2 is OFF) and then the structure is left to freely oscillate during a time (while the excitation switch 1 is OFF).

The electrostatic excitation is connected between two metal electrodes ( and ) and the silicon resonator, as illustrated in Fig. 6. These electrodes are placed diagonally as shown in the layout of Figs. 3 and 4. By using the electrodes in this configuration one can select a balanced torsion high-Q vibration mode for the tested resonators while other (unwanted) vibration modes are suppressed [6], [7]. The excitation voltage is generated by a square wave generator (“oscillator 2” in Fig. 6). Then the signal is amplified in a voltage-boosting step via a transistor and transferred to the excitation electrodes where it alternates between 0 and 150 V. C. Detection The capacitance variations due to the oscillation are sensed between silicon and two metal electrodes placed diagonally ( and or and ), as shown in the layout of Figs. 3 and 4. The vibrations of the resonator are detected by a MOSFET-transistor. The capacitor placed immediately before the MOSFETtransistor is used to suppress the bias voltage superimposed with the oscillating detected signal (the nominal cut-off frequency of the RC highpass filter before the MOSFET is approximately 1 kHz). The latter is transferred at the MOSFET-transistor input, then amplified in two amplification steps and fed to a counter which displays the measured frequency. When the same electrodes are used for both excitation and detection, an additional switch (switch 3), driven by oscillation 1, is needed during the detection interval to avoid loading the detection input which has high impedance. The electronic implementation for this particular set-up is represented in Fig. 6 in the gray area. D. Frequency Locking To find the resonance frequency of the vibrating structure, an , is first adjusted by a current controller to initial frequency, oscillator 2, with help of a variable resistor. This oscillator sends


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Fig. 6. Block diagram of the electronics. The signals A, B and C are represented in Fig. 2. The gray area shows the set-up adopted when the same electrodes were used for both excitation and detection.

a “burst” of pulses to the resonator during the excitation period. Then a detection signal comes back and is sent to a phase-detector that can detect a 90 phase-shift, meaning resonance. If no resonance is detected, the phase-shift detector output signal will vary the “voltage control” of oscillator 2, which in turn will ramp up or down its frequency; see Fig. 6. This process is repeated until resonance is found. At this point, the phase-sensitive circuit output only slightly changes the “voltage control” of oscillator 2 in order to follow and lock onto the resonance frequency. This completes the feedback control loop and maintains the resonator at resonance. The phase-shift circuitry can find and control any resonance in a range of 70% of the iniand 1.7 ). Note that tial frequency (i.e., between 0.3 the feedback control loop only occurs during the detection time . During the excitation period, the phase-sensitive circuitry is connected to a “sample & hold” circuit in order to avoid any perturbation for the phase locking during that time.

E. Charging Effect To eliminate the previously described parasitic charging effect problem at the glass surface, the capacitance formed by the resonator and the detection electrodes is alternately charged to bias detection voltages of 150 V and 150 V through a re). This sistor, at half of the frequency of oscillator 1 (i.e., is represented by switch 4 in Fig. 6. Another switch (switch 5), ), and combined with running at the same frequency ( two amplifiers of different polarity, is also introduced in the electronics, see Fig. 6. This is to always obtain an output detection voltage of identical polarity, regardless of the polarity of the bias detection voltage.

V. PRINCIPLE VERIFICATION MEASUREMENTS For the measurements presented in this section, , , , and were set to 15 Hz, 33 ms, 33 ms, 1.6 ms, and 25 ms, respectively. Note that the voltage measurements were all taken after the first amplification step at point , see Fig. 6. The excitation voltage was alternating between 0 and 150 V, while the detection bias voltage was oscillating between 150 V and 150 V. A. Silicon Resonant Density Sensor A SEM-picture of a cross-section of the low-pressure encapsulated densitometer tested is shown in Fig. 7, where the internal cavity pressure is 1 mbar [10]. When using the “burst” technique, the electrodes for excitation and detection were placed outside the internal cavity on top of the glass lid, at a distance of 400 m from the resonator. No electrical feedthoughs were needed and a large resonator to lid wall distance of 100 m could be formed reducing squeeze-film damping losses and consequently increasing the quality factor. The measured Q-value was 3400, which is of the same order of magnitude as for the same unencapsulated resonator where squeeze-film damping can be neglected [11]. The Q-factor was measured by disconnecting the feedback control and manually adjusting the resonance frequency. To measure the performance/frequency of the silicon tube densitometer, it was filled with air and the initial frequency for ) was set to 10 000 Hz. Using the feedback conoscillator 2 ( trol loop, the frequency was ramped up and resonance was found and locked at 16 513 Hz. The tube was then filled with different solutions of 2-propanol in water. The feedback control loop was able to continuously follow the variations of the resonance fre-

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Fig. 7. SEM-photo showing a cross-section of the tested density sensor. The electrodes used for excitation and detection are placed on top of the glass lid. No electrical feedthroughs are needed. The distance between the glass surface and the silicon tube is approximately 100 m. Fig. 9. Measured resonant frequencies of the pressure sensor versus applied pressure, using the “burst” technology.

B. Silicon Resonant Pressure Sensor

Fig. 8. Measured resonant frequencies of the density sensor versus liquid density for five different mixtures of water/2-propanol. The “burst” electronics with feedback-loop control enabled to make on-line measurements of liquid densities.

quency while changing the fluid density inside the tube. The resonance frequency with water in the tube was 13 006 Hz. Output ) were obtained which signals of several volts (typically 8 V is far superior to the mV levels that we obtained using standard electronics combined with internal electrodes [4]. The measured , resulting in a signal-to-noise ratio noise level was 75 mV (SNR) of 100 for the densitometer. The measured resonance frequency versus density is shown in Fig. 8. By the straight line in the figure we can see that the theoretical relation [6] is accurately measured by the new ”burst” electronics with feedback-loop control. We were also able to confirm that an excitation voltage oscillating between 0 and 15 V together with a detection bias voltage alternating between 150 V and 150 V was sufficient for frequency locking. Frequency locking was also obtained for an oscillating excitation voltage between 0 and 40 V with a detection bias voltage alternating between 40 V and 40 V. In both cases, the output voltage was about 150 mV peak-to peak, which is also the minimum amplitude to obtain frequency locking with this circuit.

Encapsulated inside a low-pressure cavity (1 mbar) the silicon resonant pressure sensor presented a Q-factor of 5400 [7]. As the densitometer, the electrodes are placed on top of the glass lid. The distances resonator-glass lid wall and resonator-electrodes are 30 m and 470 m, respectively. The “burst” technology enabled continuous measurements of the applied pressure. The resonance frequency locked by the feedback control loop was 28 042 Hz for a pressure of 1000 mbar. The measured resonance frequency versus applied pressure is shown in Fig. 9, showing the linear relation between preswere measure and frequency [7]. Output voltages of 27 V , meaning an SNR sured with a noise signal level of 75 mV as high as 360. Another test with the pressure sensor was performed to verify if the same electrodes could be used for both excitation and deand were used, and a switch (switch tection. Electrodes 3) was introduced as shown in the gray area of Fig. 6. The results showed that the same electrodes could be used for both excitation and detection with frequency locking. However, due to the output capacitance of the inserted switch ( 12 pF), the measured amplitude was reduced by a factor of approximately two, compared to measurements performed without using the same electrodes. VI. ADVANTAGES OF THE METHOD The described method has several advantages. High excitation voltages can be combined with highly sensitive detection electronics while avoiding crosstalk between excitation and detection. Elimination of “on-chip” electronics is also possible, making hybrid electronics connected to a separate sensor a fast and easy way to prototype a production instrument. Excitation and detection are separated in time meaning that the same electrodes can be used for both excitation and detection. Another advantage of the “burst” electronics is its high flexibility and adaptability: it can be applied to several types of excitation/detection techniques and to a large number of silicon resonators without needing any major design changes.


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Fig. 10. Schematic cross sections of two encapsulated resonators, (a) one with conventional small cavity depth with electrodes close to the resonator [4] and (b) one with a large gap and external electrodes using the “burst” technology.

As illustrated in Fig. 10, the “burst” electronics can also be very advantageous for encapsulated resonators. It is for instance possible to use electrodes on the outside of the resonator encapsulation. This simplifies the fabrication of these chips since no feedthroughs are needed and leads to less risk of leakage (better long term stability) and increased yield. Since more sensitive electronics can be used, larger resonator-electrode distances can be utilized and thus higher Q-values can be obtained. For instance, by increasing the distance resonator-lid wall from 30 to 100 m the Q-value increases from 3000 [resonator of Fig. 10(a)] to 9000 [resonator of Fig. 10(b)].

Fig. 11. Illustration of the rotating wheel equipment setup used to determine the resolution of the “burst” detection electronics.

VII. ELECTRONIC EVALUATION A. Power Consumption The “burst” electronics circuit had a supply voltage of 15 V and the measured power consumption was 4 W, independent of the level of the excitation voltage. Note that the electronics was not optimized to obtain the smallest possible power consumption. To reduce it, the present electronic components might be replaced by fewer and low-power consumption components. Miniaturization of the electronics using an ASIC circuit might further reduce the power consumption. B. Resolution The resolution was measured using the set-up shown in mm) was placed Fig. 11. A planar electrode (area mm at a distance from a rotating wheel (rotation frequency of 56 Hz) with 18 conductors. The distance corresponds to a capacitance calculated as follows:

(1) is the overlapping area between the fixed electrode where mm), is the vacuum permitand one conductor ( mm is the relative permitivity of air ( ). This tivity, and

M

Fig. 12. Measured peak-to-peak voltage at point (see Fig. 6) versus the capacitance formed between an electrode and rotating conductors as described in the set-up of Fig. 11.

measurement set-up can be seen as a simulation of capacitance variations during capacitive detection. The fixed electrode and the conductor formed the detection capacitance connected to the detection part of the “burst” electronics. By varying the distance between the electrode and the conductor (with help of a micrometer screw), the amplitude of (see Fig. 6) was measured. the measured signal at point The results obtained are reported in Fig. 12. We deduced that the minimum measurable capacitance was 0.4 fF, which corresponds to a peak-to-peak output voltage of approximately 1 V. , although adThe measured noise level was about 75 mV ditional optimization, such as better shielding, possibly could reduce it further resulting in higher sensitivities. However, this does not mean that 0.4 fF is the smallest capacitance change that can be detected with this circuit. With a peak voltage of 1 V for 0.4 fF and a noise level of only 75 mV, one could expect that the circuit is able to detect a 0.04 fF change.

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Fig. 13. Measured peak-to-peak voltage at point (see Fig. 6) versus the number of excitation periods per cycle, for the tested pressure sensor. These measurements were performed for different duty cycles and “burst” frequencies ). Note that two periods at a duty cycle of 1% and at a “burst” frequency ( of 115 Hz were sufficient for exciting the resonator and locking the resonance frequency (28 042 Hz).

f

C. “Burst” Cycle As explained in the previous sections, the feedback control loop used to maintain the resonator at resonance only occurs during the detection time . This means that during the excitation time , the time delay and the time left after the de), tection just before a new excitation period (i.e., no feedback control is applied. These “blind” time windows are undesirable since they can lead to a loss of resonance. This may occur if a quick frequency change, due to a change in the physical parameter one wants to measure, occurs during one of the blind time windows. These time windows should be minimized if one wants to apply a feedback control loop almost continuously. On the other hand, to keep the resonator at resonance and to select the desired resonance frequency, the excitation should be applied during enough time, and it should be repeated relatively often. Considering the above, we investigated the influence of different parameters to optimize and better understand the behavior of the “burst” electronics. One parameter was the number of excitation periods during one “burst” cycle. Another parameter was the frequency of oscillator 1 (the “burst” frequency), which dictates how often the “burst” cycle is repeated. Finally, the “burst” duty cycle, defined as the excitation time compared to ], was investigated. the time of free oscillation [i.e., A short duty cycle means a short excitation time relative to the time of free oscillation. By varying these parameters, we mea(see Fig. 6) of sured the peak-to-peak output voltage at point the resonant pressure sensor. The time delay was set to 1.6 ms and the detection time to 6.8 ms. The electronics was the limiting factor for having a shorter time delay, the structure relaxation for signal stabilization being much faster. This time delay could further be reduced by using other electronic components. The results obtained are reported in Fig. 13. For a given resonance frequency (mode of vibration) and a given duty cycle, the number of excitation periods is a function . For a given “burst” duty cycle, of the “burst” frequency a change in the number of excitation periods (or in the “burst”

frequency ) does not have a strong influence on the mea, one tends to decrease sured output voltage. By decreasing the output voltage due to a reduction in the repetition of excitadecreases (with a same duty tion cycles. However, while cycle), the number of excitation periods per cycle increases, which tends to increase the output voltage. The results obtained demonstrate that these two effects canceled out, the net result being a fairly constant output voltage. From Fig. 13, one can clearly see that the parameter with the most influence on the output detection voltage is the “burst” equal to 15 Hz, the output duty cycle. For example, with to 25 V when the duty cycle voltage increased from 1 V was increased from 1% to 50%. This voltage increase is due to a longer excitation time relative to the free oscillation time. As previously discussed, a high “burst” frequency and a short duty cycle are highly desirable. In the measurements shown in Fig. 13, we were able to lock the resonance frequency (28 042 Hz) with only two excitation periods at a duty cycle of 1% and a “burst” frequency of 115 Hz while the corresponding output voltage was 600 mV. VIII. CONCLUSIONS A method with feedback-loop control for detection and excitation of resonant silicon sensors was presented and tested for electrostatic excitation and capacitive detection. Using this method, the crosstalk between excitation and detection of silicon resonators was eliminated. This technology based on discontinuous “burst” excitation was tested on both a resonant fluid densitometer and a pressure sensor with outside glass lid electrodes. Using this technology, no electrical feedthrough conductors were needed and a large cavity gap of 100 m between the resonator and the lid wall could be formed for the density sensor resulting in a low squeeze-film damping and a high quality factor of 3400. The feedback control electronics enabled frequency locking and online measurements of liquid density and pressure with SNRs as high as 100 for the density sensor and 360 for the pressure sensor. It was demonstrated that the same electrodes could be used for both excitation and detection. An electronic resolution of 0.4 fF was measured and could be combined with high excitation voltages alternating between 0 and 150 V. The electronic evaluation revealed that the “burst” duty cycle (i.e., the excitation time relative to the free oscillation time) had a strong influence on the output detection voltage. It was also shown that only two excitation periods at a “burst” duty cycle of 1% and a “burst” frequency of 115 Hz were sufficient to select and lock the resonance frequency (28 042 Hz) for the pressure sensor. A solution was presented to solve the problem of parasitic charging effect due to localized charge storage on dielectric surfaces placed between the resonator and the detection electrodes. REFERENCES [1] J. C. Lötters et al., “Design, fabrication and characterization of a highly symmetrical capacitive triaxial accelerometer,” Sens. Actuators A, vol. 66, pp. 205–212, 1998. [2] J. Bienstman, J. Vandewalle, and R. Puers, “The autonomous impact resonator: A new operating principle for a silicon resonator strain gauge,” Sens. Actuators A, vol. 66, pp. 40–49, 1998.


[3] C. Linder, E. Zimmermann, and N. F. d. Rooij, “Capacitive polysilicon resonator with MOS detection circuit,” Sens. Actuators A, vol. 25–27, pp. 591–595, 1991. [4] T. Corman, P. Enoksson, and G. Stemme, “Low-pressure-encapsulated resonant structures with integrated electrodes for electrostatic excitation and capacitive detection,” Sens. Actuators A, vol. 66, pp. 160–166, 1998. [5] A. Fartash, I. K. Schuller, and M. Grimsditch, “Vibrating membrane elastometer for reliable measurement of mechanical properties of metallic films,” Rev. Sci. Instrum., vol. 62, pp. 494–501, 1991. [6] P. Enoksson, G. Stemme, and E. Stemme, “Fluid density sensor based on resonance vibration,” Sens. Actuators A, vol. 47, pp. 327–331, 1995. [7] J. Melin, P. Enoksson, T. Corman, and G. Stemme, “A low pressure encapsulated DRIE resonant pressure sensor electrically excited and detected using “burst” technology,” in Proc. Micromech. Eur. Conf., Gif-sur-Yvette, France, Aug.–Sept. 2, 1999, pp. 97–100. [8] M. K. Andrews, G. C. Turner, P. D. Harris, and I. M. Harris, “A resonant pressure sensor based on a squeezed film of gas,” Sens. Actuators A, vol. 36, pp. 219–226, 1993. [9] J. Wibbeler, G. Pfeifer, and M. Hietschold, “Parasitic charging of dielectric surfaces in capacitive microelectromechanical systems (MEMS),” Sens. Actuators A, vol. 71, pp. 74–80, 1998. [10] T. Corman, P. Enoksson, K. Norén, and G. Stemme, “A low-pressure encapsulated resonant fluid density sensor with feedback control electronics,” Meas. Sci. Technol., vol. 11, pp. 205–211, 2000. [11] P. Enoksson, G. Stemme, and E. Stemme, “Vibration modes of a resonant silicon tube density sensor,” J. Microelectromech. Syst., vol. 5, pp. 39–44, 1996.

Thierry Corman was born in St. Etienne, France, on April 7, 1972. He received the M.Sc. degree in electrical engineering in 1995 from the National Institute of Applied Sciences (INSA), Lyon, France, and the Licentiate of engineering and Ph.D. degrees, in 1998 and 1999, respectively, both from the Royal Institute of Technology, Stockholm, Sweden. He is a Project Manager in the microsystem research and development group, ACREO AB, Kista, Sweden. His interests include encapsulation of microsensors on the wafer scale, resonant microsensors and medical applications.

Kjell Norén was born in Stockholm, Sweden, on March 19, 1948. From 1964 to 1967, he studied electronics at Enskede Senior High School and worked as an Electrical Technician at the Research Institute of National Defence, Stockholm, Sweden. In 1969, he received the electrical engineering degree from Åsö Senior High School, Stockholm. In 1970, he joined the Instrumentation Laboratory at the Royal Institute of Technology, Stockholm, where he constructed electrical equipment for research and industrial projects, and, in 1986, he became a Research Engineer at the Image Processing Laboratory. Presently, he is a Research Engineer in the Silicon Sensor Research Group, Department of Signals, Sensors and Systems, Royal Institute of Technology, Kista, Sweden, where he is involved in the electrical and mechanical measurements of many research projects.

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Peter Enoksson was born in Lindesberg, Sweden, on April 19, 1957. He received the M.Sc. degree in engineering physics, the Licentiate of Engineering, and the Ph.D. degree, in 1986, 1995, and 1997, respectively, all from the Royal Institute of Technology, Stockholm, Sweden. In 1997, he was appointed Assistant Professor at the Silicon Sensor Research Group, Department of Signals, Sensors and Systems, Royal Institute of Technology, Kista, Sweden. His research is in the field of resonant silicon sensors and actuators, especially for fluid applications.

Jessica Melin was born in Stockholm, Sweden on May 26, 1975. She received the B.A.Sc. degree in biomedical (electronics) engineering in 1998 from Simon Fraser University, Burnaby, BC, Canada. Presently, she is a Ph.D. student in the Silicon Sensors Research Group, Department of Signals, Sensors, and Systems, Royal Institute of Technology, Stockholm. Her research focuses on microsensors and microdevices for biomedical applications with a special emphasis on intravenous pressure sensors. She has a particular interest in biomedical microsystems and micro-total analysis systems (TAS).

Göran Stemme was born in Stockholm, Sweden, on February 4, 1958. He received the M.Sc. degree in electrical engineering in 1981 and the Ph.D. degree in solid state electronics in 1987, both from the Chalmers University of Technology, Gothenburg, Sweden. In 1981, he joined the Department of Solid State Electronics, Chalmers University of Technology, Gothenburg. There, in 1990, he became an Associate Professor (Docent) heading the silicon sensor research group. In 1991, he was appointed Professor at The Royal Institute of Technology, Stockholm. He heads the Instrumentation Laboratory at the Department of Signals, Sensors and Systems. His research is devoted to sensors and actuators based on micromachining of silicon. Dr. Stemme is a subject editor for the IEEE/ASME Journal of Microelectromechnical Systems.