Three-Phase Three-Limb Coupled Inductor for Three ... - IEEE Xplore

IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 63, NO. 1, JANUARY 2016

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Three-Phase Three-Limb Coupled Inductor for Three-Phase Direct PWM AC–AC Converters Solving Commutation Problem Ashraf Ali Khan, Student Member, IEEE , Honnyong Cha, Member, IEEE, and Heung-Geun Kim, Senior Member, IEEE

Abstract—In this paper, a family of three-phase direct pulse-width modulated (PWM) ac–ac converters consisting of buck, boost, and buck–boost converters is proposed. The proposed converters have no commutation problem even if all of the switches are turned on or off simultaneously. They do not use lossy snubbers and do not sense the voltage polarity for commutation, and produce high-quality output voltage waveforms. The proposed converters allow the use of power MOSFETs and fast recovery freewheeling diodes independently. The use of power MOSFETs as active switches and freewheeling diodes with extremely fast recovery features lower the switching losses and enable us to reduce the volume of passive components by increasing switching frequency. The input (or output) ﬁlter inductor experiences twice of the converter switching frequency and thus can be designed with minimum size. To increase power density of the proposed converters, a three-phase three-limb coupled inductor is proposed. Three-phase coupled inductor integrates three separate coupled inductors of the proposed converters in one three-limb core. In comparison with separate coupled inductors, the three-phase coupled inductor has a smaller size with large currenthandling capability. Experimental results obtained for the boost-type converter show the robustness of the proposed three-phase ac–ac converters. Index Terms—Commutation problem, MOSFET, threephase ac–ac converter, three-phase three-limb coupled inductor.

I. I NTRODUCTION

M

ANY industrial applications including motor drives, power regulators, dynamic voltage restorers, and direct power conversion in wind power generators use ac–ac power converters. The use of a pulse-width modulated (PWM) rectifier

Manuscript received January 31, 2015; revised May 4, 2015 and June 8, 2015; accepted July 1, 2015. Date of publication August 11, 2015; date of current version December 9, 2015. This work was supported by the Basic Science Research Program through the National Research Foundation of Korea (NRF) funded by the Ministry of Science, ICT, and Future Planning (NRF-2013R1A2A2A01069038). A. A. Khan and H. Cha are with the School of Energy Engineering, Kyungpook National University, Daegu 702-701, Korea (e-mail: [email protected]; [email protected]). H.-G. Kim is with the Department of Electrical Engineering, Kyungpook National University, Daegu 702-701, Korea (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIE.2015.2466552

followed by a PWM voltage-source inverter (VSI) with a DC link [1]–[3], matrix converter [4]–[6], and direct (PWM) ac–ac converter [7]–[9] are the popular topologies employed for the ac–ac power converters. Among them, the direct PWM ac–ac converters are practical choice for applications that require only voltage regulation because they have such advantages as higher efficiency, smaller size, single-stage power conversion, ease of control, low harmonic current components, and low cost [10]–[13]. Direct PWM ac–ac converters include simple ac–ac converters [10]–[12], Z-source ac–ac converters [13], switched capacitor ac–ac converters [14], and resonant ac–ac converters [15]. Although, these converters have no commutation problem with ideal gate signals, delays in switch drive circuit and finite switching speed of semiconductor switches can cause overlap or dead time in the gate signals. During the overlap time the voltage source or capacitor is short-circuited, and high-current spike destroys the semiconductor switches. Similarly, during the dead time, the inductor has no path through which to keep current continuity, and high voltage spike destroys the switches. To avoid the high voltage/current spike and provide safe commutation, most of the existing direct PWM ac–ac converters either use lossy RC snubbers or sense the voltage/current polarity. Fig. 1 shows the traditional three-phase direct PWM ac–ac converters. With ideal gate signals as shown in Fig. 2(a), they have no commutation problem. However, the overlap or dead time in the gate signals can cause commutation problems and they are illustrated in Fig. 2(b) and (c) for the buck-type three-phase ac–ac converter as an example. A common method employed to address the problem of commutation has been to add bulky and lossy RC snubbers to the traditional direct PWM ac–ac converters with finite dead time. However, the dead time reduces output voltage quality and limits the obtainable output voltage. Moreover, the power loss in the resistor of an RC snubber circuit reduces the efficiency of the converters. In addition, RC snubber circuits with dead time cannot prevent the semiconductor switches from being shortcircuited in faulty conditions. To overcome the drawbacks of lossy RC snubbers soft commutation techniques have been introduced and successfully employed in [16], [17]. However, these approaches have several disadvantages. First, they have to sense the current/voltage polarity for the commutation. Second, they are unreliable with distorted input voltages especially around the zero crossing.

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Fig. 1. Conventional three-phase direct PWM ac–ac converters. (a) Boost type. (b) Buck type. (c) Buck–boost type.

Fig. 2. Commutation problem in conventional buck type three-phase ac–ac converter. (a) Ideal gate signals. (b) Dead time. (c) Overlap time.

Fig. 3. Switching cells with coupled inductor and capacitor [18].

Third, they increase control complexity and introduce additional weight to the design. From the above discussions, it is concluded that most of the existing three (or single) phase direct PWM ac–ac converters should either use bulky and lossy RC snubbers or sense the voltage/current polarity to provide safe commutation. To solve the commutation problem mentioned above, singlephase direct PWM ac–ac converters using switching cell structure are introduced in [18]. Fig. 3 illustrates the phase-leg implementation used in [18]. As shown in Fig. 3, the conventional phase-leg is replaced with switching cells (P-cell and N-cell) [18]–[24], a coupled inductor (CL), and a capacitor (C). With this structure, the commutation problem in the conventional direct PWM ac–ac converters can be eliminated without either using lossy snubbers or sensing the voltage/current polarity for safe commutation because all the switches in the converters can be short- and open-circuited without damaging switching devices.

This paper extends the switching cell structure in [18] and [24] to three-phase direct PWM ac–ac converters and three new three-phase buck, boost, and buck–boost type ac–ac converters are obtained. Similar to the single-phase converters in [18], the proposed three-phase ac–ac converters do not suffer from commutation problem. As will be shown in Section II, the proposed three-phase ac–ac converters require three coupled inductors which are somewhat bulky. In this paper, a threephase three-limb coupled inductor integrating the three coupled inductors into one is also proposed to increase power density of the proposed converter. A 1.7 kW proposed boost-type three-phase direct PWM ac–ac converter is developed. Detailed operation of the proposed converters and three-phase three-limb coupled inductor, equivalent circuits, design methodology, and experimental results are reported herein. II. P ROPOSED T HREE -P HASE D IRECT PWM AC–AC C ONVERTER TOPOLOGIES The proposed three-phase buck, boost, and buck–boost type ac–ac converters are shown in Fig. 4. The proposed converters use P- and N-type switching cells to solve the shootthrough and dead time problems. Each switching cell shown in Fig. 3 consists of one switching device and one diode connected in series [19]. The proposed converters use power MOSFETs and freewheeling diodes independently. The freewheeling diodes (D1 −D6 ) can be chosen externally with very fast recovery features and low turn-on resistance. For this reason the

KHAN et al.: COUPLED INDUCTOR FOR THREE-PHASE DIRECT PWM AC–AC CONVERTERS

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Fig. 5. Gate signal generation of the proposed three-phase ac–ac converters.

Fig. 6. Input and circulating current distributions in the coupled inductors.

The PWM gate signals of the proposed converters are shown in Fig. 5 and they are similar to the single-phase ac–ac converters in [18]. This PWM strategy increases the effective frequency of the proposed converters and produces three-level PWM bipolar voltage between P- and N-cell (see the experimental waveforms in Fig. 15). III. A NALYSIS AND O PERATION OF THE P ROPOSED T HREE -P HASE D IRECT PWM AC–AC C ONVERTERS

Fig. 4. Proposed three-phase direct PWM ac–ac converters. (a) Buck type. (b) Boost type. (c) Buck–boost type.

proposed converters can operate with high switching frequency without reverse recovery problem, which contributes to the increase in power density and efficiency of the converter. The power MOSFETs cannot be used beneficially in the traditional ac–ac converters with high switching frequency because their body diodes are associated with reverse-recovery problems [21], [25], [26]. Three separate coupled inductors CLa , CLb , and CLc are inserted between the switch arms (S1 , S4 ), (S2 , S5 ), and (S3 , S6 ), respectively. These coupled inductors serve to limit currents when the switches in each phase or all of the switches are turned on. Three capacitors Ca , Cb , and Cc are added to provide current paths when the switches in each phase or all of the switches are turned off. These capacitors also serve as filter and lossless regenerative snubber capacitors [18]. The per-phase circuit for the proposed three-phase ac–ac converters is equivalent to that of the single-phase ac–ac converters introduced in [18].

The boost-type ac–ac converter shown in Fig. 4(b) is selected and analyzed in this paper and similar analysis can be extended to the buck and buck–boost-type converters. The proposed topologies have four continuous conduction modes as shown in Fig. 5 and three-level PWM voltage waveforms between the P- and N-cells of each phase leg. Fig. 6 shows current distribution in the three coupled inductors, CLa , CLb , and CLc . Ls represents self-inductance of the coupled inductors. The coupled inductors have two distinct current components; an input inductor current and a circulating current. The average of the winding currents (ia1 − ic2 ) results in circulating currents (icma , icmb , icmc ), and the difference in the winding currents results in input inductor currents (ia , ib , ic ) [18]. In all of the operational modes of the converters, the following relationships are always valid: ix + ix2 icmx = 1 2 ix1 = icmx − i2x ix2 = icmx + i2x ix = ix2 − ix1 where x = a, b, c, for phase a, b, and c.

(1) (2) (3)

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The following analysis is conducted for D < 0.5 and a similar analysis can be carried out for D > 0.5. It is assumed the three input inductors (La , Lb , and Lc ) are equal and represented as L. A. Mode 1 [0 ∼ DTs ] Mode 1 and its per-phase equivalent circuit are shown in Fig. 7(a) and (b). During this mode, the switches (S1 −S6 ) are all turned on, and the diodes (D1 −D6 ) are all reverse biased. As a result, the input inductors are charged. The voltage and current relationships in this mode are expressed as vo vLx = vix − x (4) 2 vi − vox /2 dix = x (5) dt L dicmx vc = x. (6) dt 4Ls B. Mode 2 [DT s ∼ 0.5Ts ] Fig. 7(c) and (d) shows mode 2 and its per-phase equivalent circuit. The switches (S1 −S3 ) remain on and (S4 −S6 ) are turned off. The diodes (D1 −D3 ) are turned on due to freewheeling action and (D4 −D6 ) remain reverse biased. As a result, the input inductors are discharged to the output capacitors and input energy is transferred to the load. The voltage and current relationships are as follows: vLx = vix − vox dix vi − vox = x dt L dicmx = 0. dt

(7) (8) (9)

C. Mode 3 [0.5T s ∼ (0.5 + D)Ts ] Mode 3 and its equivalent per-phase circuit are shown in Fig. 7(e) and (f). In this mode, all of the switches are turned off and the diodes are all turned on, and the input energy is again stored in the input inductors. The voltage and current relationships are the same as in mode 1. The circulating currents are as follows: dicmx vc =− x. dt 4Ls

(10)

D. Mode 4 [(0.5 + D) ∼ Ts ] Mode 4 is the same as mode 2. By applying the flux-balance condition on the input inductors, voltage gain of the proposed boost-type three-phase ac–ac converter can be obtained as (11) and the resultant voltage gain is the same as that of the conventional boost-type ac–ac converter 1 vox = . vix 1−D

(11)

In (11), D is the duty ratio defined as the turn-on period of the switches (S4 −S6 ) during switching period. vox and vix are the output and input phase voltage of the proposed boost-type converter, respectively.

Fig. 7. Operational modes of the proposed three-phase boost type converter. (a) Mode 1. (b) Per-phase equivalent circuit of mode 1. (c) Mode 2. (d) Per-phase equivalent circuit of mode 2. (e) Mode 3. (f) Per-phase equivalent circuit of mode 3.


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Equation (16) shows that the dc-offset flux [the first term in (16)] is proportional to the maximum input current and increases as the output power increases. If Ls is large, then the high-frequency current components are relatively small, and the flux due to the dc-offset circulating current is the main flux that needs to be considered when designing the core. To handle the dc-offset flux the separate coupled inductors must have enough airgap. B. Three-Phase Three-Limb Coupled Inductor

Fig. 8. (a) Separate coupled inductor. (b) Reluctance model of separate coupled inductor. (c) Three-phase three-limb coupled inductor. (d) Reluctance model of three-phase three-limb coupled inductor.

IV. C OUPLED I NDUCTOR S TRUCTURE FOR THE P ROPOSED T HREE -P HASE AC–AC C ONVERTERS A. Separate Coupled Inductors The circulating currents in the separate coupled inductors consist of dc offset currents and high-frequency current components. The dc-offset current components of each phase are ideally equal to half of the maximum input fundamental harmonic current (ix. max /2). The high-frequency current components of the circulating current are given by (20). Therefore, the circulating currents can be expressed as icmx =

ix. max + Δicmx . 2

(12)

For a balanced three-phase load, the dc-offset current components of each phase are same, while the high-frequency current components are different. The dc-offset current generates dcoffset flux and Δicmx generates high-frequency flux in the core. From (2), the winding fluxes (Φx1 , Φx2 ) of the three separate coupled inductors shown in Fig. 8(a) can be obtained as Ls ix. max ix + Δicmx − Φx 1 = (13) N 2 2 Ls ix. max ix + Δicmx + (14) Φx 2 = N 2 2 where N is the number of turns of each coil of the coupled inductor. The magnetic core structure of the separate coupled inductor is shown in Fig. 8(a) and (b) shows its magnetic circuit. g and c are the reluctances of the airgap and core, respectively. Fig. 8(b) shows that the sum of the winding fluxes (Φx1 , Φx2 ) gives the total flux Φx through CLx and can be obtained as Φx = Φx 1 + Φx 2 . Substituting (13) and (14) into (15) gives 2Ls ix. max Φx = + Δicmx . N 2

(15)

(16)

The magnetically independent separate coupled inductors are used at the expense of size and dc-offset fluxes. A simple way of reducing the size of reactive components is to increase the switching frequency. However, the converter efficiency will be decreased due to increased switching loss. To decrease the volume of magnetic components and increase power density of the proposed converters, the three separate coupled inductors are integrated into one three-limb core. The principles for integrating magnetic components are explained in [27], and based on these principles the integrated magnetic components are used in [28]–[30] for the purpose of magnetic volume reduction. All the windings of the separate coupled inductors in the proposed converters are magnetically coupled as shown in Fig. 8(c) and the reluctance model of the three-phase coupled inductor is shown in Fig. 8(d). The dc-offset fluxes oppose each other and cancel out completely. Only the high-frequency flux components present in the three-limb core. To simplify the analysis, all three legs are assumed to be identical and the coupled inductor is assumed to have perfect coupling. However, the coupling coefficient of coupled inductor (or transformer) cannot be unity in practice. It is obvious that the leakage inductance (Llk ) makes the voltage and current waveforms in the coupled inductor deviate from its ideal waveforms, so it should be minimized. In this paper, tight bifilar winding is used and Llk can be reduced significantly (less than 1 uH). It should be noted here that the Llk in the proposed topology is seen in series with the boost input inductor. Thus, the Llk can contribute to the reduction of the input boost inductor more or less. From Fig. 8(d), it can be observed that the fluxes produced by each leg (Φa , Φb and Φc ) split equally between the remaining two legs. Hence, the resultant fluxes Φar , Φbr , and Φcr through legs a, b, and c, respectively, can be expressed as ⎧ Ls ⎪ ⎨Φar = N (2Δicma − Δicmb − Δicmc ) (17) Φbr = LNs (2Δicmb − Δicma − Δicmc ) ⎪ ⎩ Ls Φcr = N (2Δicmc − Δicma − Δicmc ) . Equation (17) shows that the dc fluxes in each leg are completely cancelled out and only high-frequency fluxes are present in the core. The high-frequency flux cannot be removed. The proposed three-phase coupled inductor exhibits perfect dc-flux cancellation and requires no airgap if all of its legs are identical. However, small airgap should be used to handle the remaining dc flux present in the core because of mismatches among the legs and unbalanced load. In addition the use of small airgap makes the inductance of each leg identical, predictable, and

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stable. The main features of the three-phase three-limb coupled inductor are as follows: 1) High Magnetizing Inductance: The three-phase coupled inductor can achieve very high self-inductance because of very small or negligible airgap. If the dc flux is eliminated, then unlike the separate coupled inductors as given by (16), the high self-inductance of the three-phase coupled inductor does not affect the dc flux in the core. 2) Large Current Handling Capability: The perfectly designed three-phase coupled inductor has very high current handling capability because the dc fluxes produced by the circulating current are removed. This unique property makes the three-phase three-limb coupled inductor ideal for highpower three-phase ac–ac conversion applications. 3) Small Inductor Size: The three-phase coupled inductor has a unique feature in that it has large current handling capability and it requires no or very small airgap. Therefore, it can be designed with minimum number of turns and smaller core to obtain the same winding current ripples and saturation current as the separate coupled inductors. Moreover, the threephase coupled inductor requires one three-limb core to realize the magnetic integration of all of the coils of three separate coupled inductors. V. C URRENT R IPPLE A NALYSIS OF I NDUCTORS A. Input Inductor Design By combining (5) and (11), input current ripples Δix of the boost-type three-phase ac–ac converter can be expressed as Δix =

(0.5D − D2 ) vox Ts L

D(1 − D) vox Ts . Lx.conv

(19)

The maximum input current ripples in the conventional three-phase ac–ac converters occur for D = 0.5 and maximum output voltage. The input inductors of the proposed boost-type converter experience twice the converter switching frequency. Therefore, they can be designed with much smaller size than those of the conventional boost-type ac–ac converter. B. Separate Coupled Inductor Design In this paper, the leakage inductance of the separate and three-phase coupled inductors is ignored because all of the windings are tightly bifilar wound. Using (6), the circulating current ripples can be expressed as Δicmx =

DTs vc . 4Ls x

The phase under consideration has maximum circulating current ripples when one of the remaining phase leg capacitors (Ca , Cb , and Cc ) has zero voltage. C. Three-Phase Three-Limb Coupled Inductor Design The design consideration for the three-phase three-limb coupled inductor is the same as that of the separate coupled inductors. The dc flux in the separate coupled inductors can cause core saturation as the output current increases. Therefore, the airgap in these inductors should be enlarged accordingly, which decreases the inductance and increases current ripples. The dc fluxes in three-phase three limb coupled inductor are removed and there is no need to increase the airgap with increasing output current. VI. M AGNETIC VOLUME R EDUCTION

(18)

where Ts is the switching period, and L is the inductance of the input inductor Lx . The maximum input current ripples of the proposed boost-type converter occur when D = 0.25 and maximum output voltage. On the other hand, input current ripples Δix.conv of the conventional boost-type ac–ac converter having inductance Lx.conv and duty cycle D can be expressed as Δix.conv =

Fig. 9. (a) Separate coupled inductor. (b) Three-phase three-limb coupled inductor.

(20)

Fig. 9 shows core structures for the separate and three-limb coupled inductors and the detailed dimensions are Y = 2.6 cm, Y1 = 2.2 cm and X = 2.24 cm. The width (X) of the core window is kept fixed. Thus, the core window is adjusted by changing the height (Y ). As shown in Fig. 9(a), two windings for the separate coupled inductor are placed on each leg of the core. The Y value is determined by N . Consider that both the separate and three-limb coupled inductors use the same N . For the three-phase coupled inductor both the primary and secondary windings of each separate coupled inductor are wound on each limb of three-limb core as shown in Fig. 9(b). Therefore, the height of window for the three-phase inductor is increased to 2Y from Y . Referring to Fig. 9, the magnetic volumes are obtained as V1ϕ ≈ 3Ac × [2 × (2Y1 + Y ) + 2X]

(21)

V3ϕ ≈ Ac × [3 × (2Y1 + 2Y ) + 4X]

(22)

where V1ϕ is the total magnetic volume of three separate coupled inductors, V3ϕ is the magnetic volume of three-phase coupled inductor, and Ac is the core cross-sectional area. Using (21), (22) and the core dimensions, the minimum reduction in magnetic volume with the three-limb inductor is 32%. However, the three-limb inductor requires small or negligible airgap. Therefore, the magnetic volume can be further reduced for the same current ripples. For the same self-inductance of 400 μH


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Fig. 10. DM and CM currents distribution.

and Ac , N for the separate and three-limb inductor are 30 and 17, respectively. Thus the number of turns of two coils for the three-limb inductor are reduced from 2N = 2 × 30 = 60 to 2 × 17 = 34, therefore, the window height is decreased from 2Y = 5.2 cm to 2 × 1.47 = 2.95 cm. As a result, the reduction is increased to 44.3%. VII. EMI A NALYSIS OF THE P ROPOSED C ONVERTERS In power converters some currents termed as common mode (CM) and differential mode (DM) current flow in common and differential mode paths, respectively as shown in Fig. 10. In three phase systems, the CM current of each phase finds paths through the stray capacitors and returns through the ground path which produces CM EMI. As an example, a stray capacitor (Cg ) of 1 nF is considered between the star point of load and ground for the proposed and traditional converters as shown in Fig. 10. The stray capacitor provides path for the resultant CM current (iCM ) which can be obtained as sum of all phase currents iCM = ioa + iob + ioc .

(23)

In three phase converters the DM current (iDM.x ) of each phase returns through the other two phases and satisfy the following relation: iDM.a + iDM.b + iDM.c = 0.

(24)

If iCM is supposed to be equally and symmetrically distributed in all three phases then the phase current in terms of CM and DM currents can be expressed as [31] iox = iDM.x +

iCM . 3

(25)

It is difficult to separate the DM and CM currents and EMI in each phase [32], however if the three-phase system is considered linear, symmetrical, and time invariant, then using orthogonal transformation [33], [34], the DM and CM currents can be expressed as

2 1 iDM.a = (26) ioa − (iob + ioc ) 3 2 iCM.a =

1 iCM (ioa + iob + ioc ) = . 3 3

(27)

Fig. 11. Simulated frequency spectrum of the input currents. (a) Input current of the traditional converter. (b) Input current of the proposed converter.

To keep the CM and DM noise within limit, the CM and DM chokes (filters) are employed in power converters. It is shown in [35] and discussed in [36] that the increased effective frequency for interleaved multichannel converters make the corner frequency of both DM and CM filters higher and therefore smaller size EMI filters can be obtained and power density can be improved. The PWM modulation strategy affects the CM and DM currents and EMI. Due to different PWM strategy and structure of the proposed converters, the DM and CM currents are analyzed as discussed below. A. DM and CM EMI Fig. 11 shows the simulated frequency spectrum of input and output currents. Fig. 12 shows the simulated frequency spectrum of the CM and DM currents given by (23) and (26) for the proposed and traditional boost type ac–ac converters. The simulations are carried out for the same electrical specifications as considered in the experiment. The dead time in PWM gate signals of the traditional converter is not considered and it is well known that the dead time increases the harmonics and decreases the quality of output waveforms. Because the effective frequency for the input (or output) filter inductor of the proposed converters becomes twice of the converter switching frequency, the odd order harmonics of the input, CM and DM currents are all eliminated or negligible as shown in Figs. 11 and 12. It can also be seen from Figs. 11 and 12 that the even order harmonics of input, DM and CM currents of the proposed and traditional converters are comparable. For different range of frequencies the design of EMI filter (Both DM and CM filters) varies with the order of harmonics as described for power factor correction (PFC) converters in [35]. For PFC converters the EMI standard regulation begins at 150 kHz [35], [36]. The following two cases can be considered for the design of EMI filters for the proposed converters.

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TABLE I E LECTRICAL S PECIFICATIONS

Fig. 12. Simulated frequency spectrum of the common mode (CM) and differential mode (DM) currents. (a) DM current of the traditional converter. (b) DM current of the proposed converter. (c) CM current of the traditional converter. (d) CM current of the proposed converter.

1) Frequency Range With DM and CM-EMI Filter Design Based on Odd Order Harmonics: For this case the proposed converters can obtain significant reduction in size of DM and CM-EMI filters because all odd order harmonics of DM and CM current are eliminated or negligible, therefore the design of EMI filters is shifted to even order harmonics which have smaller magnitude and high corner frequency. The EMI filters for high corner frequency are designed with small values of inductors and capacitors. For example, consider the same case as PFC converters in which for frequency ranging from 50 kHz to 75 kHz the DM and CM-EMI filters are designed on the basis of third-order harmonic [35]. Therefore, the EMI filters for the traditional three-phase ac–ac converter are designed according to the third-order harmonic with corner frequency of 150 kHz. As shown in Fig. 12(b), for the proposed converters the thirdorder harmonic of the DM current is cancelled out and that of CM current shown in Fig. 12(d) is negligible; thus the EMI

filters design is based on fourth-order harmonic. The magnitude of fourth-order harmonic of the proposed converter is smaller than that of third-order harmonic of the traditional converter. In addition, the corner frequency of fourth-order harmonic is 200 kHz. Therefore, the size of both DM and CM-EMI filters for the proposed converters can be reduced. 2) Frequency Range With DM and CM EMI Filter Design Based on Even Order Harmonics: For this case the reduction in size of DM and CM-EMI filters is not achieved because all even order harmonics are unchanged when compared to the even order harmonics of the traditional converters, however the odd order harmonics are eliminated for the proposed converter. VIII. E XPERIMENTAL R ESULTS Based on the previous analysis 1.7 kW boost-type threephase ac–ac converter is fabricated. The electrical specifications of the prototype converter and coupled inductors are shown in Table I. In this paper, several experiments are conducted at different load and power factor. A. Balanced Resistive Load The prototype converter is tested using the separate coupled inductors and three-phase three-limb coupled inductor, respectively. Both types of coupled inductor are bifilar wound. As a proof of concept, the three-phase coupled inductor is fabricated using I cores. The self-inductances for both types of inductors are designed to be the same (400 uH). Although the airgap in


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Fig. 13. Waveforms of three-phase boost type ac–ac converter with separate coupled inductors (vo = 220 Vrms, Po = 1.7 kW and D = 0.4). (a) Input voltages. (b) Output voltages. (c) Input currents. (d) Output currents.

Fig. 14. Waveforms of three-phase boost type ac–ac converter with three-phase three limb coupled inductor (vo = 220 Vrms, Po = 1.7 kW and D = 0.4). (a) Input voltages. (b) Output voltages. (c) Input currents. (d) Output currents.

the three-limb inductor can be eliminated ideally, small airgap (0.5 mm) is inserted in each leg to handle the remaining dc fluxes caused by the mismatches among legs and the unbalanced load. The separate coupled inductors require 2.2 mm airgap and comparatively large number of turns to obtain the same self-inductance as the three-phase coupled inductor. Experimental results for Y-connected resistive load (Ra = Rb = Rc = 28 Ω) are shown in Figs. 13–16. The experimental results for both types of inductors are compared in Figs. 13–16. Figs. 13 and 15 show the experimental results with the separate coupled inductors. Figs. 14 and 16 are the experimental results with the three-phase three-limb coupled inductor. Figs. 15(b) and 16(b) show that all switches can be simultaneously turned on and off without generating high voltage/current spikes and the proposed converters produce three-level bipolar output PWM voltage waveforms. From the waveforms in Figs. 13 and 14, it is found that all the waveforms in the proposed converters are clean and there is no significant distortion in the waveforms.

Figs. 15(a) and 16(a) show the drain-to-source voltage waveforms of switches and the voltage stress of the coupled inductor in phase a. Figs. 15(c) and 16(c) show current waveforms of the input and coupled inductor. It can be seen from Figs. 15(d) and 16(d) that the input inductors experience twice of the converter switching frequency. The experimental waveforms of the separate and three-phase coupled inductor are identical except the waveforms shown in Figs. 15(c) and 16(c), and it can be seen from these figures that the current ripples of the threephase coupled inductor are higher for the same self-inductance because of flux cancellation similarly as for interleaved inverter in [30]. Since the three-phase coupled inductor has the same self-inductance for lesser turns therefore its current ripples can be further decreased by increasing its number of turns. The current ripples shown in Figs. 15(d) and 16(d) are comparable when the number of turns of each coil of the three-phase inductor is increased by 6 turns and the obtained magnetic volume reduction is 38.5%. Experimentally, it is proved that the three-phase inductor is capable of handling very large current.

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Fig. 15. Waveforms of three-phase boost type ac–ac converter with separate coupled inductors (vo = 220 Vrms, Po = 1.7 kW and D = 0.4). (a) Voltage stresses of switches and coupled inductor. (b) Expanded waveforms of (a). (c) Waveforms of input current and winding currents. (d) Expanded waveforms of (c).

Fig. 16. Waveforms of three-phase boost type ac–ac converter with three-phase coupled inductor (vo = 220 Vrms, Po = 1.7 kW and D = 0.4). (a) Voltage stresses of switches and coupled inductor. (b) Expanded waveforms of (a). (c) Waveforms of input current and winding currents. (d) Waveforms of input current and winding currents for N = 23.

B. Highly Distorted Input Voltage The proposed converters can operate properly even with significantly distorted input voltages. Fig. 17 shows the waveforms with highly distorted input voltages. As shown, there is no noticeable overshoot in the switch voltage waveforms. C. Inductive Load To prove operation of the proposed converter with different power factor, Y-connected inductive load (R = 5 Ω, L = 2.5 mH) is used. Due to unavailability of large value inductor in the laboratory, load resistance is reduced instead. Thus, the converter is not operated with the rated output voltage. Fig. 18(a) shows the experimental waveforms with power factor of 0.98 (= cos 11◦ ). Fig. 18(b) shows the simulation results with rated output voltage and 0.727 power factor (lagging).

Fig. 17. Waveforms with highly distorted input voltages. (a) Voltage stress of switch, input current, and output voltage. (b) Zoom in waveforms of (a).


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Fig. 20. Waveforms with highly distorted input voltages and partially inductive unbalanced load.

Fig. 18. Waveforms with partially inductive load. (a) Experimental waveforms. (b) Simulated waveforms.

Fig. 21. Measured efficiency with balanced resistive load.

Finally, waveforms with highly distorted input voltages and partially inductive unbalanced load are shown in Fig. 20 for Za = 25 + 0.53j, Zb = 30 + 0.94j, Zc = 20 + 0.94j, where Za , Zb and Zc are the load impedances of phase a, b and c, respectively. The measured efficiency of the proposed boost-type threephase ac–ac converter as a function of duty ratio (D) is plotted in Fig. 21. The slight difference in the efficiency is mainly caused by the different current ripples and number of turns in both types of inductors. IX. C ONCLUSION

Fig. 19. Waveforms with unbalanced resistive load. (a) Input and output voltage waveforms. (b) Output currents. (c) Voltage stresses of switches.

D. Unbalanced Load Fig. 19 shows the waveforms of the proposed converter with partially unbalanced Y-connected resistive load (Ra = 20 Ω, Rb = 25 Ω, Rc = 30 Ω). As shown, the output voltages are well balanced, but the output current waveforms are not balanced and the phase with Ra = 20 Ω is carrying higher current.

In this paper, a family of highly reliable direct PWM threephase ac–ac converters has been presented. Compared to the conventional counterparts, the proposed converters have the following merits: • safe commutation without either adding lossy snubbers or sensing current/voltage polarity; • proper operation even with a highly distorted input voltage; • small-size filter inductors; • high-frequency operation and no reverse recovery problem of body diode; • maximum obtainable output voltage; • high quality output waveforms due to the absence of lossy snubbers and multilevel PWM output voltage; • more immune to EMI noise.

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Although three additional separate coupled inductors are required for the proposed converters, the volume of separate coupled inductors is reduced significantly by integrating all of their windings into one three-limb core. As a result, overall volume of the magnetic components of the proposed converters can be comparable to that of the conventional ac–ac converters. Detailed experimental results obtained for the boost-type threephase ac–ac converter using separate and three-phase coupled inductor verify the analysis and performance of the proposed three-phase ac–ac converters.

R EFERENCES [1] J. Kolar, T. Friedli, J. Rodriguez, and P. Wheeler, “Review of three-phase PWM AC–AC converter topologies,” IEEE Trans. Ind. Electron., vol. 58, no. 11, pp. 4988–5006, Nov. 2011. [2] E. C. d. Santos, C. B. Jacobina, N. Rocha, and E. R. C. da Silva, “Six-phase machine drive system with reversible parallel AC–DC–AC converters,” IEEE Trans. Ind. Electron., vol. 58, no. 5, pp. 2049–2053, May 2011. [3] D. C. Lee and Y. S. Kim, “Control of single-phase-to-three-phase AC/DC/AC PWM converters for induction motor drives,” IEEE Trans. Ind. Electron., vol. 54, no. 2, pp. 797–804, Apr. 2007. [4] E. Karaman, M. Farasat, and A. M. Trzynadlowski, “A comparative study of series and cascaded Z-source matrix converters,” IEEE Trans. Ind. Electron., vol. 61, no. 10, pp. 5164–5173, Oct. 2014. [5] S. Liu, B. Ge, X. Jiang, H. A. Rub, and F. Z. Peng, “Comparative evaluation of three Z-source/quasi-Z-source indirect matrix converters,” IEEE Trans. Ind. Electron., vol. 62, no. 2, pp. 692–701, Feb. 2015. [6] M. Hamouda, H. F. Blanchette, and K. A. Haddad, “Indirect matrix converters’ enhanced commutation method,” IEEE Trans. Ind. Electron., vol. 62, no. 2, pp. 671–679, Feb. 2015. [7] S. Jothibasu and M. K. Mishra, “An improved direct AC–AC converter for voltage sag mitigation,” IEEE Trans. Ind. Electron., vol. 62, no. 1, pp. 21–29, Jan. 2015. [8] S. M. Ahmed et al., “Simple carrier-based PWM technique for a three-tonine-phase direct AC–AC converter,” IEEE Trans. Ind. Electron., vol. 58, no. 11, pp. 5014–5023, Nov. 2011. [9] B. H. Kwon, G. Y. Jeong, S. H. Han, and D. H. Lee, “Novel line conditioner with voltage up/down capability,” IEEE Trans. Ind. Appl., vol. 49, no. 5, pp. 1110–1119, Sep./Oct. 2002. [10] S. Srinivasan and G. Venkataramanan, “Design of a versatile three-phase AC line conditioner,” in Proc. IEEE Ind. Appl. Soc. Annu. Meeting, Oct. 1995, vol. 3, pp. 2492–2499. [11] Z. Fedyczak, R. Strzelecki, and G. Benysek, “Single-phase PWM AC/AC semiconductor transformer topologies and applications,” in Proc. 33rd Annu. IEEE Power Electron. Spec. Conf., Jun. 2002, pp. 1048–1053. [12] F. Z. Peng, L. Chen, and F. Zhang, “Simple topologies of PWM AC–AC converters,” IEEE Power Electron. Lett., vol. 1, no. 1, pp. 10–13, Mar. 2003. [13] Y. Tang, S. Xie, and C. Zhang, “Z-source AC–AC converters solving commutation problem,” IEEE Trans. Power Electron., vol. 22, no. 6, pp. 2146–2154, Nov. 2007. [14] T. B. Lazzarin, R. L. Anderson, and I. Barbi, “A switched-capacitor threephase AC–AC converter,” IEEE Trans. Ind. Electron., vol. 62, no. 2, pp. 735–745, Feb. 2015 [15] H. Sarnago, O. Lucia, A. Mediano, and J. Burdio, “Efficient and costeffective ZCS direct ac–ac resonant converter for induction heating,” IEEE Trans. Ind. Electron, vol. 61, no. 5, pp. 2546–2555, May 2014. [16] B. H. Kwon, B. D. Min, and J. H. Kim, “Novel commutation technique of AC–AC converters,” Proc. Inst. Elect. Eng.—Elect. Power Appl., vol. 145, no. 4, pp. 295–300, Jul. 1998. [17] J. H. Kim, B. D. Min, B. H. Kwon, and S. C. Won, “A PWM buck–boost AC chopper solving the commutation problem,” IEEE Trans. Ind. Electron., vol. 45, no. 5, pp. 832–835, Oct. 1998. [18] H. Shin, H. Cha, H. Kim, and D. Yoo, “Novel single-phase PWM AC–AC converters solving commutation problem using switching cell structure and coupled inductor,” IEEE Trans. Power Electron, vol. 30, no. 4, pp. 2137–2147, Apr. 2015. [19] L. M. Tolbert, F. Z. Peng, F. H. Khan, and S. Li, “Switching cells and their implications for power electronic circuits,” in Proc. IEEE IPEMC, 2009, pp. 773–779.

[20] C. Liu et al., “Cascaded dual-boost/buck active-front-end converter for intelligent universal transformer,” IEEE Trans. Ind. Electron., vol. 59, no. 12, pp. 4671–4680, Dec. 2012. [21] P. Sun, C. Liu, J.-S. Lai, C.-L. Chen, and N. Kees, “Three-phase dualbuck inverter with unified pulse width modulation,” IEEE Trans. Power Electron., vol. 27, no. 3, pp. 1159–1167, Mar. 2012. [22] X. Zhang and C. Gong, “Dual-buck half-bridge voltage balancer,” IEEE Trans. Ind. Electron., vol. 60, no. 8, pp. 3157–3164, Aug. 2013. [23] Z. Yao, L. Xiao, and Y. Yan, “Dual-buck full-bridge inverter with hysteresis current control,” IEEE Trans. Ind. Electron., vol. 56, no. 8, pp. 3153– 3160, Aug. 2009. [24] A. A. Khan, H.-H. Shin, H. Cha, and H.-G. Kim, “Novel three-phase PWM AC–AC converters solving commutation problem,” in Proc. IEEE IPEC, May 2014, pp. 110–116. [25] L. Saro, K. Dierberger, and R. Redl, “High-voltage MOSFET behavior in soft-switching converters: Analysis and reliability improvements,” in Proc. 20th IEEE Telecommun. Energy Conf., 1998, pp. 30–40. [26] X. D. Huang, H. J. Yu, J.-S. Lai, A. R. Hefner, and D. W. Berning, “Characterization of paralleled super junction MOSFET devices under hard and soft-switching conditions,” in Proc. 32nd IEEE Power Electron. Spec. Conf., 2001, vol. 4, pp. 2145–2150. [27] P. Zumel, O. Garcia, J. A. Cobos, and J. Uceda, “Magnetic integration for interleaved converters,” in Proc. IEEE Appl. Power Electron. Conf., 2003, vol. 2, pp. 113–1149. [28] D. Pan, X. Ruan, C. Bao, W. Li, and X. Wang, “Magnetic integration of the LCL filter in grid-connected inverters,” IEEE Power Electron. Lett., vol. 29, no. 4, pp. 1573–1578, Apr. 2014. [29] J. Ewanchuk and J. Salmon, “Three-limb coupled inductor operation for paralleled multi-level three-phase voltage sourced inverters,” IEEE Trans. Ind. Electron., vol. 60, no. 5, pp. 1979–1988, May 2013. [30] A. M. Knight, J. Ewanchuk, and J. C. Salmon, “Coupled three-phase inductors for interleaved inverter switching,” IEEE Trans. Magn., vol. 44, no. 11, pp. 4119–4122, Nov. 2008. [31] M. Hartmann, H. Ertl, and J. W. Kolar, “EMI filter design for a 1 MHz, 10 kW three-phase/level PWM rectifier,” IEEE Trans. Power Electron., vol. 26, no. 4, pp. 1192–1204, Apr. 2011. [32] F. D. Torre, S. Leva, and A. P. Morando, “A physical decomposition of three-phase variables into common and differential mode quantities,” in Proc. IEEE Int. Symp. Electromagn. Compat., Munich, Germany, Sep. 24–28, 2007, pp. 127–130. [33] A. Consoli, G. Oriti, A. Testa, and A. L. Julian, “Induction motor modeling for common mode and differential mode emission evaluation,” in Proc. IEEE IAS Annu. Meeting, Oct. 6–10, 1996, vol. 1, pp. 595–599. [34] R. Zhang, X. Wu, and T. Wang, “Analysis of common mode EMI for three-phase voltage source converters,” in Proc. 34th IEEE Power Electron. Spec. Conf., Jun. 15–19, 2003, vol. 4, pp. 1510–1515. [35] C. Wang, M. Xu, F. C. Lee, and B. Lu, “EMI study for the interleaved multi-channel PFC,” in Proc. IEEE Power Electron. Spec. Conf., 2007, pp. 1336–1342. [36] P. Zumel, O. Garcia, J. A. Oliver, and J. A. Cobos, “Differential-mode EMI reduction in a multiphase DCM flyback converter,” IEEE Trans. Power Electron., vol. 24, no. 8, pp. 2013–2020, Aug. 2009.

Ashraf Ali Khan (S’15) was born in 1989 in Pakistan. He received the B.E. degree in electronics engineering from National University of Sciences and Technology (NUST), Pakistan, in 2012. He is currently working toward the M.S. degree leading to the Ph.D. degree in the School of Energy Engineering, Kyungpook National University, Daegu, Korea. His current research interests include magnetics, dc–dc converters, buck–boost inverters, solid-state transformers, and ac–ac converters. Mr. Khan has received many scholarships and awards for his excellent academic performance. He also received the 3rd Best Paper Award at ECCE ASIA 2015.


Honnyong Cha (M’10) received the B.S. and M.S. degrees in electronics engineering from Kyungpook National University, Daegu, Korea, in 1999 and 2001, respectively, and the Ph.D. degree in electrical engineering from Michigan State University, East Lansing, MI, USA, in 2009. From 2001 to 2003, he was a Research Engineer with the Power System Technology (PSTEK) Company, An-san, Korea. From 2010 to 2011, he worked as a Senior Researcher at the Korea Electrotechnology Research Institute (KERI), Changwon, Korea. In 2011, he joined Kyungpook National University as an Assistant Professor in the School of Energy Engineering. His current research interests include high-power dc–dc converters, dc–ac inverters, Z-source inverters, and power conversion for electric vehicles and wind power generation.

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Heung-Geun Kim (SM’12) was born in Korea in 1956. He received the B.S., M.S., and Ph.D. degrees in electrical engineering from Seoul National University, Seoul, Korea, in 1980, 1982 and 1988, respectively. Since 1984, he has been with the Department of Electrical Engineering, Kyungpook National University, Daegu, Korea, where he is currently a Full Professor and the Director of the Microgrid Research Center. He was a Visiting Scholar with the Department of Electrical and Computer Engineering, University of Wisconsin–Madison, Madison, WI, USA, from 1990 to 1991, and with the Department of Electrical Engineering, Michigan State University, East Lansing, MI, USA, from 2006 to 2007. His current research interests are ac machine control, PV power generation, and micro-grid systems.