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This is the author's version of an article that has been published in this journal. Changes were made to this version by the publisher prior to publication. The final version of record is available at http://dx.doi.org/10.1109/TIE.2014.2345336

IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS

1

Y-Source Boost DC/DC Converter for Distributed Generation Yam P. Siwakoti, Poh Chiang Loh, Frede Blaabjerg, Søren Juhl Andreasen and Graham E. Town Abstract—This paper introduces a versatile Y-source boost DC/DC converter intended for distributed power generation, where high gain is often demanded. The proposed converter uses a Y-source impedance network realized with a tightly coupled three-winding inductor for high voltage boosting that is presently unmatched by existing impedance networks. The proposed converter also has more variables for tuning the required gain, and hence more degrees of freedom for meeting design constraints. These capabilities have been demonstrated by mathematical derivation and experimental testing. For the experiments, a 300-W prototype has been built in the laboratory using Silicon Carbide devices for better efficiency. The prototype has been tested with a regulated power supply, before operating it with a high temperature Proton Exchange Membrane (PEM) fuel cell. Results obtained confirm the practicality and performance of the proposed converter. Index Terms — Y-source, Z-source, impedance networks, DC/DC converters, fuel cells, distributed generation.

D

I. INTRODUCTION

istributed Generation (DG) powered by green energy sources like solar, wind and fuel cells has gradually been recognized as a reliable and low cost way of providing premium power for meeting future energy demand. Realizing such a green distributed grid however requires more demanding power electronic converters for boosting the low source voltages to a predefined grid voltage. Consider fuel cells for example, their power densities are comparably high, which together with their low emission and quiet operation, make them attractive as distributed sources [1]. However, their DC terminal voltages are usually low, and vary widely. To illustrate, [1] and [2] are referred to, where it is stated that the terminal voltages of fuel cells are usually lesser than 50V for units rated from 5 to 10kW. The limit is higher at 350V for units rated above 300kW, which however, might not be plentiful as compared to the smaller units. Converters with high voltage gains are thus needed in general for tying fuel cells to the distributed grid. Their ranges of gain variation must also be wide for tracking the wide output voltage variations of fuel cells, which are usually reflected by their static voltage versus current (V-I) characteristics. Manuscript received October 4, 2013; revised February 1, 2014, April 4, 2014, and May 29, 2014; accepted July 10, 2014. Copyright © 2014 IEEE. Personal use of this material is permitted. However, permission to use this material for any other purposes must be obtained from the IEEE by sending a request to [email protected]. Yam P. Siwakoti and Graham E. Town are with the Department of Engineering, Macquarie University, NSW 2109, Australia (e-mail: [email protected]; [email protected]). Poh Chiang Loh, Frede Blaabjerg and Søren Juhl Andreasen are with Department of Energy Technology, Aalborg University, Aalborg 9220, Denmark (e-mail: [email protected]; [email protected]; [email protected]).

The same low voltage limitation and variation are also experienced by other green sources like solar and small wind turbines. Being renewable too, performances of these sources can further be complicated by weather and other environmental concerns [3]-[5], which undeniably, cannot be controlled. It is hence important to design high gain converters for these sources, whose gains should ideally nullify the source variations before a regulated voltage can be obtained for grid interfacing. The converters designed should also match with other requirements of the intended sources, which for solar and fuel cells, are to avoid negative currents flowing into them and limit ripple currents drawn from them [6]. The converter types can be DC/DC, AC/AC, AC/DC or DC/AC depending on the system under consideration. For DC/DC converters, conventional boost [7], push-pull [2], halfbridge [8], full-bridge [9] and those with numerous voltage multiplier cells [10] have been tried for boosting the low source voltages to a regulated DC-link voltage between 200V and 600V depending on the system requirements [11]-[14]. The amount of voltage boosting, if sizable, usually demands the inclusion of high frequency transformers in the converter circuits, whose turns ratios are almost always increased proportionally with the demanded gains. The resulting turns ratios and total number of turns might therefore be high if high gain and satisfactory coupling are to be ensured simultaneously [9] [43]. Alternatively, cascaded or multilevel techniques can be introduced to the converters for raising their gains [15] [16] [47]-[54] but the number of components and complexity involved will undoubtedly increase. Another approach is to use two-winding coupled inductors for voltage boosting, whose turns ratios can be kept comparably low even at high voltage gain. Converters implemented with coupled inductors can therefore have higher power densities, as compared to the earlier mentioned techniques. Strictly, coupled inductors are different from the high-frequency transformers mentioned earlier since their magnetizing inductances are intentionally designed to be finite (ideally infinite for transformers). Based on this approach, a few magnetically coupled impedance networks have been explored, whose common origin dated back to the Z-source impedance network introduced in [17]. The Z-source network and its subsequent quasi [18] [19], embedded [20] [21] and series [22] variations use two inductors and two capacitors for voltage boosting. They have so far been demonstrated for DC/AC inverter [17]-[23], DC/DC converter [24] [25], AC/AC converter [26] [27] and AC/DC converter [28], but their gains are lately mentioned to be low. That then leads to the development of the switched-inductor [29] [30], tappedinductor [31], cascaded [32], T-source [33] [34], trans-Zsource [35], -source [36] and TZ-source [37] networks. The latter four networks can be considered different from the others since they use magnetically coupled inductors for gain boosting. They are also different from those push-pull

Copyright (c) 2014 IEEE. Personal use is permitted. For any other purposes, permission must be obtained from the IEEE by emailing [email protected].



2 TABLE I GAINS AND DIODE VOLTAGE STRESSES OF DIFFERENT NETWORKS Diode Voltage Network Gain ̂ ⁄ Stress / Z-, Quasi-Z- and ( ) Embedded-Ẑ ⁄ source [17]-[21]

vL D1 iIn VIn

N3

N1 N2

vO

SW VC1

C1

a) D1 iIn VIn

vL N1 N2 C1

D1 N3 SW VC1

iIn VIn

Single coupled inductor

vL N1 N2 C1

N3 SW

vO

T- and Trans-ZSource [33]-[35]

[

(

)

]

-Source [36]

[

(

)

]

VC1

Y-Source

b) c) Fig. 1. Illustration of (a) Y-source impedance network, (b) its equivalent shoot-through and (c) non-shoot-through circuits.

and bridge-based converters mentioned above in the sense that their gains are increased at a faster rate than the usual proportional relationship with turns ratios. Their turns ratios and total number of turns can therefore be comparably lesser, while retaining the necessary magnetic coupling at high power density. It is hence the intention of this paper to continue with the investigation by proposing a Y-source network implemented as a high boost DC/DC converter for use with DG. The resulting converter uses a three-winding coupled inductor for flexibly deciding on its gain, which is presently not matched by related converters. The number of components used is also kept small to allow the converter to be implemented compactly without compromising its performance. The converter has been tested with a regulated DC supply before operating it with a fuel cell. Results obtained have confirmed the practicality and performance expected from the converter. II. Y-SOURCE IMPEDANCE NETWORK This section introduces the Y-source impedance network, and discusses its differences when compared with other existing impedance networks. The presented network will then be used for implementing the Y-source boost converter proposed for DG in the next section. A. Circuit Analysis The Y-source impedance network is shown in Fig. 1(a). It comprises an active switch SW, a passive diode D1, a capacitor C1 and a three-winding coupled inductor (N1, N2, N3) for introducing the high boost at a small duty ratio for SW. As the windings of the coupled inductor are connected directly to SW and D1, its coupling must be tight to ensure very small leakage inductances at its winding terminals. This can be done by following the winding style used to wind a biﬁlar coil [39] which is also the method used in [35]. However, in [35], only two wire strings are simultaneously wound to give two coupled windings, while for the Y-source inductor, three strings are wound to form three coupled windings. Upon ensuring the necessary coupling, the equivalent circuit obtained by turning on SW is shown in Fig. 1(b), where D 1 has reverse-biased naturally, and hence replaced by an opencircuit in series with the source. Expressions for this ⁄ equivalent circuit can be written as (1), where

(

[

)

̂ ⁄

̂ ⁄ ̂ ⁄

]

Single coupled inductor with an additional inductor and capacitor Improved Trans-Z[ ( ) ] ( )̂ ⁄ Source [38] Two coupled inductors (Subscript “total” represents the sum of two turns ratios) TZ-Source [37]

[

⁄

and inductor.

(

)

)̂ ⁄

(

]

are turns ratios of the three-winding coupled

⁄



⁄

.

(1)

By turning off SW subsequently, the equivalent circuit changes to Fig. 1(c), where D1 now conducts to link the source with the impedance network. This leads to expressions given in (2). 

⁄

(

).

(2)

Taking the state space average of (1) and (2) then leads to the expression in (3) for computing voltage across capacitor C1. In (3), represents the normalized on-time of SW, which classically, has been referred to as its duty ratio. (

)(



)

(

(

)⁄[

)

].

(3)

The expression in (3), when used with the equivalent circuit in Fig. 1(c), subsequently leads to the expression in (4) for computing the peak output voltage ̂ , and hence the gain ̂ ⁄ . In (4), is a term introduced to represent the winding factor of the integrated magnetics. ̂

(

̂

⁄[

) (

( )

]

)( ⁄[

)( ].

) (4)

The gain obtained from (4) and those of other classical impedance networks [33]-[38] are summarized in Table I, from which it can be seen that the gain of the Y-source network can be set higher than that of the traditional Z-source




Fig. 2. Theoretical voltage gain of the Y-source impedance network when realized with different winding factor K.

3 techniques are thus proposed in the literature [40]-[42] for obtaining a higher voltage boost at a smaller duty cycle. The amount of extra active and passive components needed is however enormous, rendering the multiplier techniques as not cost and size competitive. This problem is fortunately not experienced by the Y-source network, whose gain of ⁄( ) can conveniently be tuned high without using excessive components. The Y-source network is therefore proposed and compared with existing networks that use coupled inductors for voltage boosting in this subsection. For that, Table I summarizes gain expressions of some recently proposed impedance-source networks that use coupled-inductors, such as the T-source, trans-Z-source and source networks. To facilitate their comparison, expressions of these existing networks are rewritten as (8) and (9), where ( ) and  ( ).

. These inequalities give

[

(

rise to those winding design requirements spelled in (5) and (6) for obtaining a higher gain for the Y-source network when compared with the traditional Z-source network.

[

(

network if

and



(5)



and

.

(6)

By setting the denominator of (4) to be greater than zero, the range of variation for of the Y-source network can also be determined as (7). The range in (7) is always narrower than of the traditional Z-source network as long as the conditions in (5) and (6) are satisfied. The Y-source network is therefore capable of producing a high gain at a small duty ratio, as proven by the expressions derived. (



)

.

(7)

To further illustrate the gain performance of the Y-source network, Fig. 2 plots its gain at different values of K. As anticipated, the plot shows the different ranges of variation for the different K values considered. It also includes an example horizontal equal gain line, whose gain of 4 is always obtained ⁄ , by setting as 75% of its maximum value regardless of the actual K value chosen. Other equal gain lines can similarly be drawn, where necessary. B. Comparison with Other Coupled-Inductor-Based Networks The voltage gains of classical boost and buck-boost (including Ćuk, SEPIC and Zeta) converters are derived as ⁄( ) and ⁄( ), respectively, where D is the switch duty ratio, and hence the same as . For high gain, D is close to 1, which in practice, is not realizable since the switch is almost always turned on, and hence separating the source and load almost fully. Various voltage multiplier

]

) )

( ]

) (

(8) ) .



(9)

Parameters and  in (8) and (9) can then be related to the Y-source gain expression according to (10), where ( )( ) . [

(

)

]

[

(

)(

)

]

( ) . (10) The gain produced by the Y-source network is thus a combination of those expected from the T- or trans-Z-source and -source networks. This either creates a higher gain or allows merits of the existing networks to be flexibly merged. The latter can be helpful when choosing magnetic core, wire and coupling method for winding the three-winding inductor, which will usually face some constraints like size and type availabilities. To better illustrate this flexibility, Table II is referred to, where different winding turns ratios (N1:N2:N3) have been grouped in the last column. Each of these groups gives rise to the same winding factor K, voltage gain ̂ ⁄ and range for . Their final selection is thus decided by other physical requirements and constraints related to coupling and availability of components. Besides the T-source, trans-Z-source and -source networks, expressions for the TZ-source network in [37] and improved trans-Z-source network in [38] are also added to Table I. Strictly, these networks are extended from the transZ-source network by adding either extra passive components or a second two-winding coupled inductor. They should hence be grouped differently like shown in Table I. The same extensions and derivations, where necessary, can also be applied to the Y-source network by adding the same extra passive components or a second three-winding inductor [32] [44]. These extensions are however not discussed further since the preference here is to keep the component count low, while not compromising gain. More comparative details of these impedance-source networks (and others) can be found in [45], [46], where the attractiveness of the Y-source network has



IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS further been strengthened in terms of voltage gain, component count and power density. In addition to the impedance-source networks, networks with either a single or multiple coupled-inductors included, but with different operating principles, have also been proposed in [47]-[54]. The aim is still to increase their overall gains, whose expressions, device stresses and component counts are summarized in Table III for comparison with the proposed Y-source network. It is evident from the table that voltage gains of the summarized existing networks are linearly raised by increasing their coupled turns ratios n (in the numerators). In contrast, gain of the Y-source network is exponentially raised by increasing its winding factor K (in the denominator). The total number of turns needed by the Ysource network for producing a higher voltage gain is thus TABLE II GAIN OF Y-SOURCE IMPEDANCE NETWORK WHEN REALIZED WITH DIFFERENT WINDING FACTOR K AND TURNS RATIO (N1:N2:N3)

(

)

3

(

)

4

(

)

5

(

)

6

(

)

7

(

)

8

(

)

9

(

)

10

(

)

1:1:3, 2:1:4, 1:2:5, 3:1:5, 4:1:6, 1:3:7 1:1:2, 3:1:3, 2:2:4, 1:3:5, 4:2:5 2:1:2, 1:2:3, 5:1:3, 4:2:4, 8:1:4 3:1:2, 2:2:3, 1:3:4, 7:1:3, 6:2:4 4:1:2, 3:2:3, 2:3:4, 1:4:5, 9:1:3 5:1:2, 4:2:3, 3:3:4, 2:4:5 6:1:2, 5:2:3, 4:3:4, 3:4:5, 2:5:6 7:1:2, 6:2:3, 5:3:4, 4:4:5 8:1:2, 7:2:3, 6:3:4

TABLE III COMPARISON OF VARIOUS COUPLED-INDUCTOR-BASED DC-DC CONVERTERS Number of Components Voltage** Device Gain Reference Coupled Stress*** Switch Diode Capacitor ( ) Inductor [49]

1

3

1 with 2 windings

4

[50]

2

1

1 with 2 windings

2

1

2

1 with 2 windings

2

[52]

1

6

1 with 2 windings

6

[53]

5

0

1 with 2 windings

3

1

2

1 with 2 windings

2

1

2

1 with 3 windings

2

[51]

[55]*

Proposed Y-source

(

(

)

)

a

D1 iIn

c

vL N3

N1

Note: The above gain and stress expressions are for ideal cases with perfect coupling. * Where coupled Ćuk, SEPIC and Zeta converters are collectively studied. ** Where ( ) and n is the coupled magnetic turns ratio. *** Where Vin and Vo are the converter input and output voltages, respectively.

D2 RLoad

N2

VIn

SW b

C1

VOut

vO C2

VC1

Fig. 3. Y-source boost DC/DC converter. dstTs

Vgs

(1-dst)Ts

t

Vd1

t

Iin=Id1

t

Vds

t

Ids

t

Vc1

Gain Gv 2

4

t

Vd2

t

Id2 t0 t1 t2 Ts

t3

t

Fig. 4. Expected waveforms from the Y-source boost DC/DC converter.

comparably lesser, resulting in smaller size, cost and losses. In addition, like deduced from Table II, the same number of winding turns for the Y-source network, when orientated differently, flexibly gives rise to different gains. This can be seen by referring to the example of 3:1:5, 1:3:5 and 5:1:3, whose total turns remain unchanged, but have different voltage gains expressed as [ ] , [ ] and [ ] , respectively. This flexibility is presently not matched by existing networks using coupled inductors. III. PROPOSED BOOST DC/DC CONVERTER With its high boost ability, the Y-source impedance network is suitable for implementing high gain converters like the simple high boost DC/DC converter proposed in this paper. Principle of the proposed converter and its associated waveforms are discussed first before addressing a few tradeoffs that are also experienced by other coupled-inductorbased converters. A. Principle of Operation The proposed Y-source boost DC/DC converter is shown in Fig. 3, where only one controllable switch has been used. Also noticed is the addition of an extra diode D2 and an output capacitor C2 when compared with the basic Y-source network shown in Fig. 1(a). With these added components, turning SW on causes D1 and D2 to reverse-bias simultaneously. That leaves only C1 to charge the magnetizing inductance of the coupled inductor, and C2 to power the load. Turning SW off, on the other hand, causes D1 to conduct, and the input source to recharge C1. The input source, together with the coupled inductor, also supplies energy to recharge C2 and power the load, but only if voltage across C2 is lower than ̂ . When that happens, D2 conducts, hence linking C2 and the load to the rest of the circuit. The cycle repeats when SW turns on again. By periodically switching SW, the load voltage




5

across C2 can hence be regulated at ̂ , which accordingly to (4), represents a gain of

⁄

[

(

)

] . No

doubt, this is the maximum gain that the Y-source network can provide.

d’ST

dST

Vgs

Vd1 (K-1)VIn (1-KdST)

B. Expected Operating Waveforms Based on the described operating principle, Fig. 4 shows some waveforms expected from the Y-source boost DC/DC converter in response to the applied gate voltage to switch ( )⁄ ), its SW. When SW is on from to ( drain-source voltage collapses to zero, while its current rises to a finite value. That causes diodes D1 and D2 to become reverse-biased, and their voltages and to rise. The input current , which is also the current through D1, then falls to zero during this interval. When SW subsequently turns off at , across it rises sharply, together with and through the two diodes. These currents, in turn, charge C1 and C2 with voltage across C2 rising slightly above ̂ at . When that happens, D2 stops conducting with returned to zero. Since comes from the dc source, its reduction to zero also causes to drop. The converter remains in this state until SW turns on again at ( ), causing the waveforms to repeat their earlier patterns. For and , it should also be mentioned that their patterns can change slightly depending on the charging time constant of C2. A longer charging time will no doubt lengthen the interval between and , while a shorter charging time will shorten it. This variation will however not affect the maximum gain that can be produced by the converter. C. Semiconductor Voltage Stresses Like those coupled-inductor-based converters proposed in [33]-[38], semiconductor devices of the Y-source boost DC/DC converter experience instantaneous voltage stresses, whose values must be considered when selecting components for realization. Beginning with SW and D2, their stresses ̂ and ̂ can be determined by analyzing the shoot-through and non-shoot-through states, respectively. The expressions obtained are summarized in (11), where it can clearly be seen that both devices must block the same peak output voltage. ̂

⁄[

̂

].

(11)

The same expression in (11) is however not applicable to D1, whose expression should instead be determined from those preliminary expressions found in (12) for representing the shoot-through state. ̂

⁄

̂

.

(12)

Substituting (3) and the definition of K into (12) then leads to (13) for computing the peak voltage stress across D1. ̂

(

)

⁄[

].

(13)

) times larger, meaning Comparing with (11), (13) is ( D1 is stressed more even though for only a short shoot-through

(K’-1)VIn (1-K’d’ST )

t0

t1

t2

t3

Ts

Fig. 5. Illustration of instantaneous voltage stress experienced by diode D1.

duration. This is, in fact, the case for all coupled-inductorbased converters proposed in [33]-[37]. To explain how the high instantaneous voltage stress is produced, the average voltage across D1 in Fig. 1(a) should first be determined as , since average voltages across the inductor windings are zero. Alternatively, the average voltage across ( ) D1 can be determined as ̂ , which when combined with the former, leads to ̂ ( )⁄ . From (3), the understanding drawn is that a certain , and hence value, can be obtained from different combinations of K and . For cases with high K and small , their instantaneous ̂ values will thus be high even if their average and values remain unchanged. This is better illustrated by the two examples shown in Fig. 5, whose averages have been kept constant. The case represented by solid lines has been confined to a shorter shoot-through time, whose instantaneous ̂ value is thus higher to keep the integration area unchanged over a switching period. It is therefore important to consider all design constraints and availabitity of components thorougly, before choosing the right combination of K and for realizing the proposed Ysource boost converter. IV. EXPERIMENTAL RESULTS To demonstrate its efficacy, a 300-W Y-source boost DC/DC converter has been built in the laboratory using components and parameters listed in Table IV. A picture showing the implemented converter can be found in Fig. 6. For the coupled inductor, its windings were realized with 80, 16 and 48 turns, whose normalized turns ratio is then given as 5:1:3. These windings can, in total, be placed in six different ways with different winding factors K produced in turn. Three possibilities are given in Table V, and drawn in Fig. 7 to appropriately show the required dot conventions. This flexibility is not found with other coupled-inductor-based converters proposed in [33]-[38], which use only two windings with only two different ways of placement. Out of the two possibilities, only one option gives rise to a raised boost effect. The proposed Y-source converter is therefore more versatile, and for its three winding combinations shown in Table V, it should also be mentioned that their accompanied shoot-through durations have been fixed at 75% of the maximum possible values , unless stated otherwise. According to Section-II(A), that causes the converter gain to remain at 4 even though its winding factor K changes from 2




6

to 4. The case of K = 2 should also be mentioned as similar to the traditional Z-source network according to (3) to (7). It can hence be treated as the base case for comparison.

A. Powered by Regulated DC Source Beginning with K = 4, and a dc source of 60V powering the Y-source converter, Fig. 8(a) shows the observed input and output waveforms, captured with a 500MHz digital scope from Yogogawa (DL9040). The output voltage is read as 240V, which indeed matches the expected gain of 4. Other waveforms measured across the semiconductor devices are shown in Fig. 8(b). These waveforms again match the theoretical expectations with voltages across the devices dropping to zero when they are conducting, and rising to finite values when they are blocking. It should also be noted that the conduction time of D 2 is ) , and hence shorter than that of D1. shorter than ( As explained in Section III(B), this is expected with the conduction time of D2 decided by the charging time constant of C2. Measurements were next captured with the same K = 4, but with other parameters changed to and V to produce roughly the same output voltage of 242.5V read from Fig. 9(a). The gain obtained is thus higher at 6, which again matches expectation.

Fig. 6. Experimental Y-source boost DC/DC converter. TABLE IV TESTED PARAMETERS Y- Source Converter Parameter/Description Value/Part Number Power Rating 300W 60V Input Voltage 240V Output Voltage Capacitances C1 & C2 470µF @ 400V Kemet Magnetic Core Magnetics C055863A2 MPP core Switching Frequency fs 12.6kHz C2M0080120D Silicon Carbide Power Switch SW MOSFET Z-FET™ MOSFET Diode D1 C3D25170H Silicon Carbide Schottky Diode Z-Rec™ Rectifier C3D20060D Silicon Carbide Schottky Diode D2 Diode Z-Rec™ Rectifier Serenus 166 AIR C V2.5 PEM Fuel Cell Number of Stacks 1 Number of Cells / Stack 65 Active Cell Area 45cm2 Nominal Power 1kW Nominal Voltage 31.5V Nominal Current 32A Idle Voltage 50V (can surge to 65V)

a

n

Winding Factor K

dST = 75%  dST,max

Gain

2 3 4

0.375 0.25 0.185

4 4 4

1

80 turns

2

1

16 turns

2

1

48 turns

b

2

a

c

1

80 turns

2

1

16 turns

2

1

48 turns

2

5:1:3 K=4

n

a

80 turns

2

1

16 turns

2

1

48 turns

2

VOut, 50V/div

IOut, 1A/div

vgs, 10V/div vds, 100V/div

c n

vd1, 200V/div

b

1:3:5 K=3

1

iIn, 5A/div

(a)

TABLE V TESTED WINDING PARAMETERS Turns Ratio N1: N2: N3 3:1:5 1:3:5 5:1:3

VIn, 20V/div

vd2, 100V/div

c

b

3:1:5 K=2

Fig. 7. Winding arrangements for different K values (see corresponding a, b and c terminals in Fig. 3).

(b) Fig. 8. Experimental (a) input voltage VIn, input current iIn, output voltage VOut, and output current IOut, (b) gate-source voltage of SW vgs, drain-source voltage of SW vds, voltage across D1 vd1, and voltage across D2 vd2 of Y-source converter with , K = 4 and .




7

VIn, 20V/div vgs, 10V/div iIn, 5A/div

VOut, 50V/div

IOut, 1A/div

vds, 100V/div

vd1, 200V/div

vd2, 100V/div

(a) (b) Fig. 9. Experimental (a) input voltage VIn, input current iIn, output voltage VOut, and output current IOut, (b) gate-source voltage of SW vgs, drain-source voltage of SW vds, voltage across D1 vd1, and voltage across D2 vd2 of Y-source converter with , K = 4 and .

VIn, 20V/div

vgs, 5V/div

iIn, 5A/div

vds, 100V/div

VOut, 50V/div vd1, 200V/div IOut, 1A/div

vd2, 100V/div

(a) (b) Fig. 10. Experimental (a) input voltage VIn, input current iIn, output voltage VOut, and output current IOut, (b) gate-source voltage of SW vgs, drain-source voltage of SW vds, voltage across D1 vd1, and voltage across D2 vd2 of Y-source converter with , K = 3 and .

VIn, 20V/div

vgs, 5V/div

iIn, 5A/div

vds, 100V/div

VOut, 50V/div

vd1, 200V/div

IOut, 1A/div vd2, 50V/div (a) (b) Fig. 11. Experimental (a) input voltage VIn, input current iIn, output voltage VOut, and output current IOut, (b) gate-source voltage of SW vgs, drain-source voltage of SW vds, voltage across D1 vd1, and voltage across D2 vd2 of Y-source converter with , K = 2 and .




8

Similar waveforms for K = 3 and K = 2 are also shown in Fig. 10 and Fig. 11, respectively, for a common gain of 4 (fixed by setting ) and an input voltage of 60V. The output voltage values read from Fig. 10(a) and Fig. 11(a) are hence similar, and equal to 240V, as anticipated. The actual shoot-through durations in Fig. 10(b) and Fig. 11(b) are also appropriately lengthened to ⁄ ⁄ and , as intended, to arrive at the common gain of 4. Efficiencies of the converter under different operating conditions are next measured and plotted in two different ways, as shown in Fig. 12. Fig. 12(a), in particular, is for the case of keeping the gain at 4 even with different K values used. Fig. 12(b), on the other hand, is for a fixed K of 4, whose corresponding gain will hence increase with . For an overview of the converter loss distribution, a thermal image of the air-cooled experimental prototype operating at rated 300W in the steady state was captured and shown in Fig. 13. The warmest device is identified as switch SW, which conducts and commutates the high shoot-through current. The high shoot-through current also flows through capacitor C1, and windings N2 and N3 of the coupled inductor, but their temperatures are comparably lower than that of SW. Main concerns of the Y-source converter have thus been identified as the high shoot-through current that flows through SW and the high voltage pulses observed across diode D1 in the earlier experimental figures. These concerns are however not new to the proposed Y-source converter. They are experienced by all existing converters implemented with two-winding coupled inductors.

B. Powered by PEM Fuel Cell The proposed Y-source converter was next tested with a high temperature PEM fuel cell system, whose characteristics and pictorial illustration can be found in Table IV and Fig. 14, respectively. The system was operated with ambient air supplied by its blower and clean pressurized hydrogen fuel supplied and regulated by a Greenlight Innovation fuel cell test station. The system temperature was controlled by superpositioning an additional cooling flow to the cathode air supply, and was ensured by the integrated fuel cell controller. The static V-I characteristic of the fuel cell is shown in Fig. 15, where its voltage is observed to drop as the load current and power increase at normal ambient temperature. The amount of variation can be more than a factor of 2 from noload to full-load conditions [17], which is certainly not appropriate for grid interfacing. To obtain a better regulated voltage, the Y-source boost converter was thus added after the fuel cell, and to load the converter, a 100-W electronic load was connected to its output.

Pure H2 Supply Serenus 166 AIR C V2.5

Converter

Fig. 14. Y-source converter powered by PEM fuel cell system.

(a)

(b)

Fig. 12. Efficiency of Y-source converter with varying (a) K and (b) D1 (55 0C)

.

SW (72.5 0C) D2 (40 0C)

Coupled Inductor (45 0C)

C2 (30 0C)

C1 (50 0C)

Fig. 13. Infrared image of Y-source boost DC/DC converter operating at rated 300W.

Fig. 15. Polarization curve of Serenus 166 Air C fuel cell.

Fig. 16(a) shows the obtained results, where the fuel cell voltage, which is also the input voltage of the converter, is read as 37.5V at 2.67A and 120C. With K = 4 and , the converter gain is 4, which means its output should be 150V. This is indeed the value read from the third trace in




VIn, 10V/div

9 high voltage pulses across its input diode. These devices are therefore at higher risks, and must hence be sized appropriately. Upon ensuring that, anticipated gain performance of the converter has been tested in experiments with a regulated dc supply and then a PEM fuel cell. Results obtained are in agreement with expectation, hence confirming the flexibility and suitability of the converter for high gain applications such as in distributed generation.

iIn, 1A/div

REFERENCES [1] VOut, 50V/div [2] IOut, 0.5A/div (a)

[3] [4] [5]

vgs, 5V/div [6] vds, 50V/div

[7]

vd1, 200V/div [8] [9]

vd2, 50V/div (b) Fig. 16. Experimental (a) input voltage VIn, input current iIn, output voltage VOut, and output current IOut, (b) gate-source voltage of SW vgs, drain-source voltage of SW vds, voltage across D1 vd1, and voltage across D2 vd2 of Y-source converter with K = 4 and (supplied by fuel cell).

[10]

Fig. 16(a) at a measured efficiency of 92%. Other voltage waveforms measured across the semiconductor devices are shown in Fig. 16(b) with the same expected pulsing behaviors observed. The earlier discussed operating principle and anticipated waveforms have hence been validated appropriately, demonstrating the efficacy of the converter.

[12]

[11]

[13]

[14]

V. CONCLUSION A high gain DC/DC converter operating with a small switch duty ratio has been proposed in the paper. The proposed converter uses a unique Y-source impedance network built with a three-winding coupled inductor. Turns ratio and winding placement of this inductor can be designed to give the desired gain, while maintaining a small switch duty ratio. The resulting converter is thus versatile with more degrees of freedom for tuning its gain, as compared to existing coupled-inductor-based converters. Like existing coupledinductor-based converters though, the proposed converter experiences high shoot-through current stress in its switch and

[15] [16] [17] [18] [19]

S. N. Motapon, L. A. Dessaint and K. Al-Haddad, “A Comperative Study of Energy Management Scheme for a Fuel-Cell Hybrid Emergency Power System of More-Electric Aircraft,” IEEE Trans. Ind. Electron., vol. 61, no. 3, pp. 1320-1334, Mar. 2014. G. K. Andersen, C. Klumpner, S. Kjaer and F. Blaabjerg, “A New Power Converter for Fuel Cells with High System Efficiency,” Int. J. Electron., vol. 90, no. 11/12, pp. 727-750, Nov. 2003. F. Blaabjerg, Z. Chen and S. B. Kjaer, “Power Electronics as Efficient Interface in Dispersed Power Generation Systems,” IEEE Trans. Power Electron., vol. 19, no. 5, pp. 1184-1194, Sep. 2004. Y. P. Hsieh, J. F. Chen, T. J. Liang and L. S. Yang, “Novel High Stepup DC-DC Converter for Distributed Generation System,” IEEE Trans. Ind. Electron., vol. 60, no. 4, pp. 1473-1482, Apr. 2013. B. Ge, H. Abu-Rub, F. Z. Peng, Q. Li, A. T. de Almeida, F. J. T. E. Ferreira, D. Sun and Y. Liu, “An Energy Stored Quasi-Z-Source Inverter for Application to Photovoltaic Power System,” IEEE Trans. Ind. Electron., vol. 60, no. 10, pp. 4468-4481, Oct. 2013. Fuel Cell Handbook, 6th Edition, DOE/NETL-2002/1179, US Dept. of Energy, pp. 8.27-8.29, Nov. 2002. B. Huang, A. Shahin, J. P. Martin, S. Pierfederici and B. Davat, “High Voltage Ratio Non-isolated DC-DC Converter for Fuel Cell Power Source Applications,” in Proc. IEEE-PESC’08, 15-19 June 2008, pp. 1277-1283. F. Z. Peng, H. Li, G. J. Su and J. S. Lawler, “A New ZVS Bidirectional DC/DC Converter for Fuel Cell and Battery Application,” IEEE Trans. Power Electron., vol. 19, no. 1, pp. 54-65, Jan. 2004. J. Wang, F. Z. Peng, J. Anderson, A. Joseph and R. Buffenbarger, “Low Cost Fuel Cell Converter System for Residential Power Generation,” IEEE Trans. Power Electron., vol. 19, no. 5, pp. 1315-1322, Sep. 2004. Y. J. A. Alcazar, D. S. Oliveria, F. L. Tofoli and R. P. Torrico-Bascope, “DC-DC Nonisolated Boost Converter Based on the Three-State Switching Cell and Voltage Multiplier Cells,” IEEE Trans. Ind. Electron., vol. 60, no. 10, pp. 4438-4449, Oct. 2013. M. H. Todorovic, L. Palma and P. N. Enjeti, “Design of a Wide Input Range DC-DC Converter with a Robust Power Control Scheme Suitable for Fuel Cell Power Conversion,” IEEE Trans. Ind. Electron., vol. 55, no. 3, pp. 1247-1255, Jan. 2011. M. Nymand and M. A. E. Andersen, “High Efficiency Isolated Boost DC-DC Converter for High Power Low Voltage Fuel Cell Applications,” IEEE Trans. Ind. Electron., vol. 57, no. 2, pp. 505-514, Feb. 2010. X. Liu, H. Li and Z. Wang, “A Fuel Cell Power Conditioning System With Low-Frequency Ripple-Free Input Current Using a ControlOriented Power Pulsation Decoupling Strategy,” IEEE Trans. Power Electron., vol. 29, no. 1, pp. 159-169, Jan. 2014. A. K. Rathore and Prasanna U R, “Analysis, Design, and Experimental Results of Novel Snubberless Bidirectional Naturally Clamped ZCS/ZVS Current-Fed Half-Bridge DC/DC Converter for Fuel Cell Vehicles,” IEEE Trans. Ind. Electron., vol. 60, no. 10, pp. 4482-4491, Oct. 2013. W. Li, X. Lw, Y. Deng, J. Liu and X. He, “A review of Non-isolated High Step-Up DC/DC Converters in Renewable Energy Applications,” in Proc. IEEE-APEC’09, pp. 364-369. W. Li and X. He, “Review of Nonisolated High-Step-Up DC/DC Converters in Photovoltaic Grid-Connected Applications,” IEEE Trans. Ind. Electron., vol. 58, no. 4, pp. 1239-1250, Apr. 2011. F. Z. Peng, “Z-Source Inverter,” IEEE Trans. Ind. Applicat., vol. 39, no. 2, pp. 504-510, Mar/Apr. 2003. J. Anderson and F. Z. Peng, “Four Quasi-Z-Source Inverters,” in Proc. IEEE-PESC’08, Jun. 2008, pp. 2743-2749, Rhodes, Greece. J. Anderson and F. Z. Peng, “A Class of Quasi-Z-Source Inverters,” in Proc. IEEE-IAS’08, Oct. 2008, pp. 1-7, Edmonton, Alberta, Canada.



IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS [20] P. C. Loh, F. Gao, F. Blaabjerg and A. L. Goh, “Buck-Boost Impedance Networks,” in Proc. EPE’07, Sep. 2007, pp. 1-10, Aalborg, Denmark. [21] P. C. Loh, F. Gao and F. Blaabjerg, “Embedded EZ-Source Inverters,” IEEE Trans. Ind. Applicat., vol. 46, no. 1, pp. 256-267, Jan/Feb. 2010. [22] Y. Tang, S. Xie and C. Zhang, “An Improved Z-Source Inverter,” IEEE Trans. Power Electron., vol. 26, no. 12, pp. 3865-3868, Dec. 2011. [23] Y. Li, S. Jiang, J. G. Cintron-Rivera and F. Z. Peng, “Modelling and Control of Quasi-Z-Source Inverter for Distributed Generation Applications,” IEEE Trans. Ind. Electron., vol. 60, no. 4, pp. 1532-1541, Apr. 2013. [24] H. Cha, F. Z. Peng and D. W. Yoo, “Distributed Impedance Network (Znetwork) DC/DC Converter,” IEEE Trans. Power Electron., vol. 25, no. 11, pp. 2722-2733, Nov. 2010. [25] Dong Cao and F. Z. Peng, "A Family of Z-source and Quasi-Z-source DC-DC Converters," in Proc. IEEE-APEC’09, pp. 1097-1101, 15-19 Feb. 2009. [26] X. P. Fang, Z. M. Qian and F. Z. Peng, “Single-Phase Z-Source PWM AC/AC Converters,” IEEE Power Electron. Letter, vol. 3, no. 4, pp. 121-124, Dec. 2005. [27] Y. Tang, S. Xie and C. Zhang, “Z-Source AC-AC Converters solving Commutation Problem,” IEEE Trans. Power Electron., vol. 22, no. 6, pp. 2146-2154, Nov. 2007. [28] K. You and M. F. Rahman, “A Matrix–Z-Source Converter with AC/DC Bidirectional Power Flow for an Integrated Starter Alternator System,” IEEE Trans. Ind. Applicat., vol. 45, no. 1, pp. 239-248, Jan/Feb. 2009. [29] M. K. Nguyen, Y. C. Lim and G. B. Cho, “Switched-Inductor Quasi-ZSource Inverter,” IEEE Trans. Power Electron., vol. 26, no. 11, pp. 3183-3191, Nov. 2011. [30] M. Zhu, K. Yu and F. L. Luo, “Switched Inductor Z-Source Inverter,” IEEE Trans. Power Electron., vol. 25, no. 8, pp. 2150-2158, Aug. 2010. [31] D. Li, P. C. Loh, M. Zhu, F. Gao and F. Blaabjerg, “Enhanced-Boost ZSource Inverters with Alternate-Cascaded Switched- and TappedInductor Cells,” IEEE Trans. Ind. Electron., vol. 60, no. 9, pp. 35673578, Sep. 2013. [32] D. Li, F. Gao, P. C. Loh, M. Zhu and F. Blaabjerg, “Hybrid-source Impedance Networks: Layouts and Generalized Cascading Concepts,” IEEE Trans. Power Electron., vol. 26, no. 7, pp. 2028-2040, Jul. 2011. [33] R. Strzelecki, M. Adamowicz, N. Strzelecka and W. Bury, “New Type T-Source Inverter,” in Proc. CPE’09, May 2009, pp. 191-195, Badajoz, Spain. [34] S. P. Kumar and P. Shailaja, “T-Shaped Z-Source Inverter,” Int. Journal Eng. Research & Tech., vol. 1, no. 9, pp. 1-6, Nov. 2012. [35] W. Qian, F. Z. Peng and H. Cha, “Trans-Z-Source Inverters,” IEEE Trans. Power Electron., vol. 26, no. 12, pp. 3453-3463, Dec. 2011. [36] P. C. Loh, D. Li and F. Blaabjerg, “-Z-Source Inverters,” IEEE Trans. Power Electron., vol. 28, no. 11, pp. 4880-4884, Nov. 2013. [37] M. K. Nguyen, Y. C. Lim and Y. G. Kim, “TZ-Source Inverters,” IEEE Trans. Ind. Electron., vol. 60, no. 12, pp. 5686-5695, Dec. 2013. [38] M. K. Nguyen, Y. C. Lim and S. J. Park, "Improved Trans-Z-Source Inverter with Continuous Input Current and Boost Inversion Capability," IEEE Trans. Power Electron., vol. 28, no. 10, pp. 4500-4510, Oct. 2013. [39] http://en.wikipedia.org/wiki/Bifilar_coil. [40] M. Prudente, L. L. Pfitscher, G. Emmendoerfer, E. F. Romaneli and R. Gules, “Voltage Multiplier Cells Applied to Non-Isolated DC-DC Converters,” IEEE Trans. Power Electron., vol. 23, no. 2, pp. 871-887, Mar. 2008. [41] J. C. R. Caro, J. C. M. Maldonado, R. S. Cabrera, A. G. Rodriguez, E. N. S. Cabrera and R. C. Ibarra, “A Family of DC-DC Multiplier Converters,” Engineering Letters, vol. 19, iss. 1, pp. 57-67, Feb. 2011. [42] F. L. Tofoli, D. de Souza Oliveira Jr., R. P. T. Bascope and Y. J. A. Alcazar, “Novel Nonisolated High-Voltage Gain DC-DC Converters Based on 3SSC and VMC,” IEEE Trans. Power Electron., vol. 27, no. 9, pp. 3897-3907, Sep. 2012. [43] Y. P. Siwakoti, P. C. Loh, F. Blaabjerg and G. E. Town, “Effects of Leakage Inductances on Magnetically-Coupled Impedance-Source Networks” in EPE-ECCE Europe 2014, 26-28 Aug. 2014 (Accepted). [44] D. Li, P. C. Loh, M. Zhu, F. Gao and F. Blaabjerg, “Generalized Multicell Switched-Inductor and Switched-Capacitor Z-Source Inverters,” IEEE Trans. Power Electron., vol. 28, no. 2, pp. 837-848, Feb. 2013. [45] Y. P. Siwakoti, F. Z. Peng, F. Blaabjerg, P. C. Loh and G. E. Town, “Impedance Source Network for Electric Power Conversion — Part I: A

10

[46]

[47] [48] [49] [50] [51] [52]

[53]

[54]

Topological Review” IEEE Trans. on Power Electron., DOI:10.1109/TPEL.2014.2313746, 2014. Y. P. Siwakoti, F. Z. Peng, F. Blaabjerg, P. C. Loh, G. E. Town and S. Yang, “Impedance Source Network for Electric Power Conversion — Part II: Review of Control and Modulation Techniques” IEEE Trans. on Power Electron., DOI:10.1109/TPEL.2014.2329859, 2014. W. Li and X. He, “Review of Nonisolated High-Step-Up DC/DC Converters in Photovoltaic Grid-Connected Applications,” IEEE Trans. Ind. Electron., vol. 58, no. 4, pp. 1239-1250, Apr. 2011. A. Tomaszuk and A. Krupa, “High Efficiency High Step-Up DC/DC Converters- A Review,” Bulletin of Polish Academy of Sciences, vol. 59, no. 4, pp. 475-483, 2011. R. J. Wai and R. Y. Duan, “High Step-Up Converter With CoupledInductor,” IEEE Trans. Power Electron., vol. 20, no. 5, pp. 1025-1035, Sep. 2005. T. F. Wu, Y. S. Lai, J. C. Hung and Y. M. Chen, “Boost Converter With Coupled Inductors and Buck-Boost Type of Active Clamp,” IEEE Trans. Ind. Electron., vol. 55, no. 1, pp. 154-162, Jan. 2008. Q. Zhao and F. C. Lee, “High-Efficiency, High Step-Up DC-DC Converters,” IEEE Trans. Power Electron., vol. 18, no. 1, pp. 65-73, Jan. 2003. Y. P. Hsieh, J. F. Chen, T. J. Liang and L. S. Yang, “Novel High StepUp DC-DC Converter With Coupled-Inductor and Switched-Capacitor Techniques,” IEEE Trans. Ind. Electron., vol. 59, no. 2, pp. 998-1007, Feb. 2012. Y. P. Hsieh, J. F. Chen, L. S. Yang, C. Y. Wu and W. S. Liu, “HighConversion-Ratio Bidirectional DC-DC Converter With Coupled Inductor,” IEEE Trans. Ind. Electron., vol. 61, no. 1, pp. 210-222, Jan 2014. Y. Berkovich and B. Axelrod, “Switched-Coupled Inductor Cell for DCDC Converters With Very Large Conversion Ratio,” IET Power Electron., vol. 4, iss. 3, pp. 309-315, 2011.

Yam P. Siwakoti (S’10) received the B.Tech. degree in electrical engineering from NIT Hamirpur, India (2001–2005), and the M.E. degree in electrical power engineering (magna cum laude) from NTNU, Norway, and Kathmandu University, Nepal, under the NOMA fellowship program (2008–2010). He received the Ph.D. degree in electronic engineering from Macquarie University, Australia, (2010–2014) under International Macquarie University Research Excellence Scholarship (iMQURES). His research interests include modeling and design of high power converter, wireless power transfer and application of new wide-band-gap semiconductor devices (GaN/SiC) for VHF power converter to improve reliability, power density and efficiency. He has published more than 22 research papers in refereed journals and conferences proceedings in the area of power electronics. He is also a frequent reviewer of the IEEE TRANSACTION ON INDUSTRIAL ELECTRONICS, the IEEE TRANSACTION ON POWER ELECTRONICS, the IEEE TRANSACTION ON INDUSTRY APPLICATION, and the IEEE TRANSACTION ON INDUSTRIAL INFORMATICS.

Poh Chiang Loh received his B.Eng (Hons) and M.Eng from the National University of Singapore in 1998 and 2000 respectively, and his Ph.D from Monash University, Australia, in 2002, all in electrical engineering. His interests are in power converters and their grid applications.




11

Frede Blaabjerg (S’86–M’88–SM’97–F’03) was with ABB-Scandia, Randers, Denmark, from 1987 to 1988. From 1988 to 1992, he received the Ph.D. degree with Aalborg University, Aalborg, Denmark. He became an Assistant Professor in 1992, an Associate Professor in 1996, and a Full Professor of power electronics and drives in 1998. His current research interests include power electronics and its applications such as in wind turbines, PV systems, reliability, harmonics and adjustable speed drives. Dr. Blaabjerg received 15 IEEE Prize Paper Awards, the IEEE PELS Distinguished Service Award in 2009, the EPE-PEMC Council Award in 2010, and the IEEE William E. Newell Power Electronics Award 2014. He was an Editor-in-Chief of the IEEE TRANSACTIONS ON POWER ELECTRONICS from 2006 to 2012. He has been Distinguished Lecturer for the IEEE Power Electronics Society from 2005 to 2007 and for the IEEE Industry Applications Society from 2010 to 2011.

Søren Juhl Andreasen was born in Aalborg, Denmark, in 1981. He received the M.Sc. degree in mechanical engineering with specialization in electromechanical system design and the Ph.D. degree from Aalborg University, in 2005 and 2009, respectively. In 2012, he was appointed Associate Professor at Aalborg University, Department of Energy Technology. He is involved in research concerning fuel cell and battery power systems involving, testing of hybrid electric systems, battery management systems, applied electrochemical impedance spectroscopy on fuel cells and lithium-ion batteries for electrical vehicle traction. His main research interests include theoretical and experimental analysis and control of fuel cell systems with a special focus on high-temperature polymer-electrolyte membrane fuel cells and methanol reformer systems.

Graham E. Town (S’87–M’89 - SM'06) received the B.E. (Hons 1) degree from the New South Wales Institute of Technology, Sydney, Australia, in 1984 and the Ph.D. degree from the University of Sydney, Sydney, Australia, in 1992. From 1978 to 1985 he was Amalgamated Wireless Australasia. In 1985, he joined the Department of Electrical Engineering, University of Sydney, to undertake PhD studies in nuclear magnetic resonance imaging, and was appointed Lecturer in 1991. From 1992 to 2000 he was also an academic member of the Australian Photonics Cooperative Research Centre. In 2002 he joined Macquarie University, Sydney, Australia, where he established the University's undergraduate engineering program, and is currently a Professor in the Department of Engineering. His research contributions have been diverse, including NMR imaging and spectroscopy, guided-wave optics and photonics, broadband and multi-wavelength fiber lasers, telecommunications regulation, radio-over-fibre systems, terahertz technology, power electronics and systems, and engineering education.
