Zero-Voltage and Zero-Current-Switching Half Bridge DC/DC Converter

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Zero-Voltage and Zero-Current-Switching Half Bridge DC/DC Converter

Abstract: This paper introduces an improved zero-voltage and zero-current-switching half bridge (ZV-ZCS HB) DC/DC converter for medium and high-power applications. The soft switching is obtained by adding only small capacitors. The operation principle and design procedure are presented together with an analysis of the total losses in the switch. By using the proposed operation principle and design procedure zero-voltagezero-current switching and high efficiency operation for full load range are provided. The operation and design considerations are illustrated and verified on a 6-kW prototype, which has been simulated (PSPICE) and successfully tested at 20 kHz operating frequency. The obtained efficiency is above 97% for full load. Increasing the operating frequency above 50kHz is possible, which contributes to a decrease of the losses in the transformer and to a reduction the size and weight. The simple topology, high efficiency and low cost make the proposed converter attractive for medium and high power applications. Keywords—Soft Switching, Zero-Voltage-Zero-Current Switching

I. INTRODUCTION Recent developments in high-frequency power conversion have shown an increased utilization of soft switching techniques. These techniques can be classified into four families [1]: resonant [2,3,4], quasi-resonant [5,6], multiresonant [7] and PWM converters with soft switching [8,9,10]. All these circuits and techniques can be regarded as resonant and soft switching PWM techniques. For resonant-type techniques the main drawbacks are significantly higher voltage and/or current stresses on semiconductors (compared with those in the conventional PMW technique) and variablefrequency control. The distinctive features of soft-switching PWM techniques are constant-frequency duty-ratio control with simultaneous reduction of switching losses for the same, or only slightly increased current and voltage stresses on the semiconductors. But those topologies usually require auxiliary passive and/or active components. Depending on the specific application, different topologies could be used according to the specific requirements. In some applications the voltage control is not needed or can be realized in another part of the total circuit. This paper presents an improved zero-voltage and

zero-current-switching half bridge (ZV-ZCS HB) DC/DC converter suitable for such conditions. A simultaneous zerocurrent and zero-voltage switching is obtained for full load range without any additional drawbacks. The converter is applicable with benefits in medium and high-power applications because of its high efficiency and optimal switching conditions. II. PRINCIPLE OF OPERATION The proposed converter (fig.1) consists of a half bridge inverter and a half bridge rectifier. The combined circuit is a high efficiency DC/DC converter with simultaneous zerovoltage and zero-current switching conditions. The features of the ZV-ZCS HB DC/DC converter have been achieved by adding only small capacitors in parallel to the switches and by a specific operation principle and components dimensioning. The zero-current turn-on and turn-off are obtained by a series resonant circuit consisting of the capacitors Cd1,Cd2 of the voltage divider and the leakage inductance of the transformer. The zero-voltage turn-on is achieved by discharging the capacitors Ck1,Ck2 with a sufficient magnetizing current. These specific operation conditions are accomplished in the proposed design procedure. An additional advantage of the circuit is its low series impedance due to the series resonance between the transformer leakage inductance and the capacitors Cd1,Cd2, resulting in a low variation of the ratio Uout /Uin. iin

iout S1

Uin

D1 Ck1

iL2

Cd1

D3 CO1

Cf

Uout iL1

S2

D2 Ck2

Cd2

D4

Co2

Fig.1. Zero-Voltage and Zero-Current-Switching Half Bridge (ZV-ZCS HB) DC/DC Converter.

Rout

Fig.2. Simulation waveforms of the proposed ZV-ZCS HB DC/DC converter (PSPICE); Uin=400V; Ck1=Ck2=16nF; Cd1=Cd2=5mF; LS1=LS2=1.25mH; Lm=500mH, Co1=Co2=500mF; Rout=25W; tON=16ms; tOFF=4ms; T=40ms; N=1. t0 – the turn-on signal is applied to S1; t1 – S1 starts conducting; t2 – the secondary winding current becomes zero; t3 – the turn-off signal is applied to S1; t4 – Ck1 is fully discharged and the voltage across S2 is zero; t5 – the turn-on signal is applied to S2; t6 – S2 starts conducting; th – the time interval during which the voltage across the switch S1 (S2) is Uin; tS – the time interval of the slope of the voltage across the switches, tS=T/2-th.

The start up is realized by a low duty ratio or a high operating frequency in order to limit the current through IGBTs during the charging of the output capacitors Cо1,Cо2. In fig.2 the simulation results (PSPICE) for the following component values are shown: Uin=400V; Ck1=Ck2=16nF;

Cd1=Cd2=5mF; LS1=LS2=1.25mH; Lm=500mH; Cо1=Cо2=500mF; Rout=25W; tON=16ms; tOFF=4ms; N=1 (N is the transformer ratio; LS1,LS2 are leakage inductances of the primary and secondary windings of the transformer; Lm is the magnetizing inductance). The IGBTs are modeled by the models of real

S1

D1

Ck1

Cd1

D3

CO1

Cd Rout

Uin

S2

D2

Ck2

Cd2

D4

LS1

LS2’

Uin

Co

Uout

Rout

Co2

Fig.3. Equivalent circuits for the time interval t1¸t2.

switches presented in [11]. The transformer ratio 1:1 is not a necessary condition, but using it allows balancing parasitic capacitances of the primary and secondary windings as well as reducing eddy currents in the windings. From fig.2 it is clear, that the half bridge inverter resembles a pure resonant inverter according to the primary winding current iL1 and it resembles a voltage inverter according the voltage across Cd1 and Cd2.

the average voltage across the primary winding is Uin/2 (th is slightly larger than tON). Then iLm(t3) is:

A) time interval t0¸t1;At the instant t0 a turn-on signal is applied to the switch S1 and the switch current becomes positive at the instant t1 after the change of the current direction in the primary winding of the transformer.

D) time interval t3¸t4; After the instant t3 a process of charging Ck1 and discharging Ck1 begins. The current through both capacitors can be considered constant because of the high value of Lm. At the end of the process Ck1 is charged to Uin and Ck2 is discharged to zero (Ck1 and Ck2 are in parallel for that process and we introduce the value Ck = Ck 1 + Ck 2 ). Using (2), the duration of the slope time interval tS=Т3,4 (tS=T/2- th) is:

B) time interval t1¸t2; At the instant t2 the current through the secondary winding is zero. The converter state during the time interval t1¸t2 is shown in fig.3. The resonant process, which begins after turning on the switch S1 includes the total leakage inductance LS (LS @ LS1+LS2’, N=1), the capacitors Cd and Co. On one hand Cd = Cd 1 + Cd 2 as C d >LS and Cd